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Differential oscilloscope probes are indispensable tools for engineers who need to measure signals accurately in complex environments. Whether you are troubleshooting everyday low-voltage circuits or tackling the challenges of high-voltage power electronics, the right probe ensures safety, precision, and reliable data capture. Yet, with so many options available—each designed for specific ranges and applications—understanding how to select and use differential probes effectively can make the difference between clear insights and misleading results.

This article explores the essentials of differential probes, highlighting their role in both common and high-voltage measurements, and offering practical guidance for engineers who want to master their use.

Understanding differential probes

At their core, differential probes are designed to measure the voltage difference between two points that are not referenced to ground. Unlike single-ended probes, which assume one side of the signal is tied to earth ground, differential probes float with the circuit under test, making them ideal for analyzing signals in isolated systems, switching power supplies, motor drives, and other environments where ground-referenced measurements can be misleading—or even unsafe.

By rejecting common-mode noise and providing accurate readings across a wide voltage range, differential probes give engineers the confidence to capture clean waveforms in both everyday low-voltage circuits and demanding high-voltage applications.

The poor man’s alternative: A-B math mode

Some engineers turn to the oscilloscope’s A–B math mode as a low-cost substitute for a true differential probe. By connecting two standard single-ended probes to separate channels and subtracting one from the other, the scope can display the voltage difference between two points. While this trick works for basic low-voltage measurements, it suffers from a critical drawback: poor common-mode rejection ratio (CMRR).

Furthermore, this method creates a dangerous grounding hazard; because standard probes remain tied to the scope’s Earth-grounded chassis, attempting this on floating high-voltage circuits can cause a catastrophic short circuit that a true, isolated differential probe would easily prevent.

Dedicated differential probes are carefully designed with matched inputs, shielding, and circuitry that reject common-mode noise and interference. In contrast, the A–B math method relies on two independent channels that rarely match perfectly in gain, phase, or frequency response.

As a result, common-mode signals leak into the measurement, producing distorted or noisy waveforms. This makes A–B math unsuitable for precision work and unsafe for high-voltage applications, where accurate rejection of common-mode voltage is essential (while floating-input oscilloscopes are an effective alternative, we will not be covering them in this post).

Figure 1 The A–B math mode on an oscilloscope uses two channels to approximate a differential measurement. Source: Author

Isolation transformers: A stopgap, not a solution

One of the most dangerous pitfalls in high-voltage oscilloscope measurements is the ground clip trap. Even if the circuit is floated, the probe’s ground clip remains internally tied to earth ground. Accidentally clipping to a high-voltage node can instantly short the circuit, destroy equipment, and pose a severe shock hazard.

A common workaround is to power the device under test (DUT) through an isolation transformer, breaking the direct connection to earth ground. This allows probes to be connected more flexibly and can make certain measurements possible when a proper probe is unavailable.

Floating a circuit also introduces new risks: exposed nodes may sit at dangerous potentials relative to ground, and the oscilloscope itself can be compromised if isolation fails. For these reasons, the 1:1 isolation transformer approach should be regarded only as a stopgap “poor man’s” option. When working with high-voltage systems, the safe and reliable solution is always a properly rated probe designed for the task.

Figure 2 A 1:1 isolation transformer lets probes connect without a ground reference, but the ground clip stays internally tied to earth and poses risk. Source: Author

It’s worth noting is that isolating the DUT—rather than the oscilloscope—is a standard power electronics practice that significantly assists a differential probe by floating the entire circuit’s reference. This setup effectively eliminates ground loops that otherwise inject EMI into your measurements via the probe’s cable shielding.

More importantly, it reduces common-mode stress on the probe’s internal amplifiers; since the DUT is no longer hard-tied to Earth ground, the probe does not have to fight a massive voltage potential relative to the scope’s chassis. This results in a much cleaner signal with higher fidelity, particularly when probing high-side MOSFETs or bridge rectifiers where the reference point is constantly swinging.

The right take: Differential scope probes

So, differential probes are specialized tools for measuring the voltage difference between two points in a circuit. They feature two inputs that can be connected anywhere without requiring a ground reference. An internal differential amplifier produces an output voltage proportional to the difference between the chosen points, typically scaled by a user-defined attenuation factor.

Figure 3 An active differential probe extends the measurement capabilities of a standard oscilloscope. Source: Pico Technology

Recall that a major advantage of differential probes is their ability to reject common-mode signals—voltages present simultaneously at both inputs. This makes them highly effective for capturing low-level signals in noisy environments. They can also be used for single-ended measurements by grounding one of the leads.

As an aside, it’s worth mentioning that a differential probe is not the same as a differential preamplifier like the Tektronix ADA400A. Probes are designed for general oscilloscope measurements across a wide bandwidth, while preamplifiers are specialized for ultra-low-level, low-frequency signals. ADA400A, for example, offers selectable gain and filtering, making it ideal for micro-volt level work in noisy environments.

Although ADA400A is still supported and available through some distributors, it’s considered more of a legacy accessory than a mainstream option. In practice, that means it remains useful for precision applications but is not promoted for new designs the way modern differential probes are. In short, use a probe for broad, everyday measurements, and reach for a preamp when chasing precision at the very bottom of the signal scale.

Getting back on track, high-voltage differential probes are among the most widely used types in modern test and measurement setups. And, galvanically isolated HV differential probes go further by providing complete electrical separation between the high-voltage circuit under test and the oscilloscope, protecting both the operator and sensitive equipment.

This isolation—often implemented through optical coupling techniques—prevents ground loops, reduces noise interference, and ensures accurate measurements even in environments with large voltage swings. Their combination of safety, fidelity, and versatility makes them indispensable tools in high-voltage and high-power applications.

As a summary (kept simple for clarity), all differential probes rely on active circuitry, since measuring the voltage difference between two points requires rejecting common-mode signals. Everyday differential active probes are used for precision work in high-speed digital and low-level analog circuits.

For power electronics, high-voltage differential active probes are the standard, enabling safe measurement of floating signals and large common-mode voltages. And when maximum safety and fidelity are needed, galvanically isolated differential probes—often using optical isolation—provide complete separation between the circuit under test and the oscilloscope, preventing ground loops and protecting both operator and equipment.

Practical session: Use cases and key specifications

This session is on the practical side, focusing on when differential probes are actually needed and the key specifications that matter most when choosing one.

Needless to say, differential probes are required whenever signals are not referenced to ground or involve large common-mode voltages. A classic case is measuring the gate-to-source voltage on a high-side MOSFET in a switching converter. Because the source terminal is floating and rides on the switching node, a standard single-ended probe tied to ground would be unsafe and misleading.

In this situation, a high-voltage differential active probe captures the true waveform safely, and if voltages or noise are extreme, an optically isolated probe adds full separation between circuit and oscilloscope for maximum protection and accuracy.

Figure 4 A practical application example using a differential probe to capture floating gate-to-source voltage signals in a power electronics circuit. Source: Author

Below are the key specifications engineers should keep in mind:

• Common mode rejection ratio (CMRR): Measures how well the probe ignores “noise” or voltages that appear equally on both leads. Note that CMRR is frequency-dependent and typically drops as the signal frequency increases. A higher CMRR ensures cleaner measurements in high-interference environments.
• Voltage rating: Defined by both differential voltage (between leads) and common-mode voltage (leads to ground), often categorized by CAT safety ratings such as CAT II and CAT III). These ratings ensure the probe can safely handle both the signal’s magnitude and any potential transients in your application.
• Attenuation ratio: Most differential probes provide fixed or switchable ratios. This setting defines how much the input signal is scaled down before reaching the oscilloscope, balancing high-voltage safety with signal fidelity.
• Bandwidth: Determines how faithfully fast signals are captured. Because square waves are composed of high-frequency harmonics, a probe’s bandwidth should ideally be 3 to 5 times higher than the signal’s fundamental frequency to avoid “rounding off” sharp transitions.
• Input Impedance: High resistance minimizes DC loading on the circuit. However, be aware that effective impedance drops significantly at high frequencies due to the effects of internal capacitance.
• Input capacitance: This is the primary factor that “slows down” fast transitions or causes circuit loading at high speeds. Lower capacitance is essential for maintaining signal integrity and preventing the probe from changing the behavior of the circuit under test.

Clearing the mist on differential probes

As often, this post also leaves some mist but hopefully clears enough to reveal the essentials. Differential probes are not exotic extras—they are the right tool whenever signals float, swing at high voltages, or demand precision beyond what a single-ended probe can safely deliver.

From active types for clean digital and analog work, to high-voltage versions for power electronics, and galvanically isolated probes for maximum safety, the choice comes down to matching probe and specs to the measurement challenge. And those specs—CMRR, bandwidth, risetime, voltage rating, attenuation ratio, input impedance, capacitance—are not just numbers; they decide whether your waveform is faithfully captured or dangerously distorted.

So next time you reach for a probe, pause to check your choice and its specs—the right differential probe is not optional, it’s essential for accuracy, safety, and confidence in your measurements.

T. K. Hareendran is a self-taught electronics enthusiast with a strong passion for innovative circuit design and hands-on technology. He develops both experimental and practical electronic projects, documenting and sharing his work to support fellow tinkerers and learners. Beyond the workbench, he dedicates time to technical writing and hardware evaluations to contribute meaningfully to the maker community.

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The post Mastering differential probes: Fundamentals and advanced insights appeared first on EDN.

I dont have more images., I used a raspberry pi pico with a voltage conversion board. the chips were taken from the disk not in a beautifull condition so I need to make these copper boards.. (actually if the chips are taken correctly there are special sockets for them). After the software was done I discovered these chips also were failing ran very hot. So it wasn't a success... submitted by /u/Distinct-Question-16 [link] [comments]

I made this atrocity with a CAN bus module, SD card module, humidity, temp, pressure, acceleration and gyro sensors. The use-case here really to extract and log everything from a CAN bus, dump it to SD and then download the data with bluetooth to an android device and push to a hosted API for analysis. Then optimize how to run an outboard engine (rpm, energy/distance, trim etc). But my point is, why didn't I do this shit 10 years ago? Or is it just that this has never been this easy before? It's just so much fun. Ignore the arduino in the background, it was my only available breadboard at the time. I'm a CS major, never really done any electronics but tons of programming on all levels. I can't understand why I have never even tried this before. The possibilities are endless! Using an ESP32-S3 Devkit for this project, which seems very capable and speaks CAN natively. Feel free to citique the soldering, it's my first time soldering small things. submitted by /u/wenoc [link] [comments]

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I love getting cool swag from hackathons and I also love designing PCB's, so when my friend asked me if I would design hackathon badges for a large game jam in singapore, I was absolutely down! The theme of overglade was a "The game jam within a game", pretty cool concept right! High schoolers from around the world were flown out to the event by hackclub after they spent about 70 hours designing their own game. These badges needed to be really cheap and simple, because we were going to manufacture about a hundred in a pretty limited amount of time. I went with a zero-power approach, which means sticking with e-inks, and I decided to include NFC if the organizers wanted to introduce it into the roleplay of the event, and so participants could add their website or github if they so choose! I used an RP2040-based architecture because it's really easy and cheap to get on the first try, and then added an ST25 passive NFC tag which was really simple to configure. The badge is in the shape of a ticket, because you got a "ticket" to the event after spending a lot of time designing games to qualify! 20 GPIO's are broken out onto the edges if you're ever in a pinch at a hackathon, and I wanted the badges to feel really fun so there's a lot of art designed by various people in the community! The badge worked really well and I learned quite a lot in the process. My takeaways are to manufacture a BUNCH of extra badges, because some will end up breaking; to think about your PCB in 3D, because one of the inductors was a bit tall and caused more badges to break; and to have a strong vision of your final product, because it really helped me to create something unique and beautiful :D The project is fully open source (https://github.com/KaiPereira/Overglade-Badges) if you want to manufacture some of your own, or reference for your own boards, and if you have any feedback or questions, I'd love to hear them! submitted by /u/KaiPereira [link] [comments]

Actually works. submitted by /u/dreaddymck [link] [comments]

My water heater recently stopped working for no obvious reason; only 2 years old. Rheem is sending me a new board to replace this one, but I thought I'd share pics of the damage for anyone interested. submitted by /u/ElectronicswithEmrys [link] [comments]

Editor’s note: This is a multi-part series on how to design a digital-controlled PFC: 

• How to design a digital-controlled PFC, Part 1
• How to design a digital-controlled PFC, Part 2
• How to design a digital-controlled PFC, Part 3

Total harmonic distortion (THD) and power factor are two major criteria used to evaluate power factor correction (PFC) performance. Meeting strict THD and power factor requirements is always a challenge for PFC designs. In this third installment of the series, I will introduce a set of digital methods to reduce THD and improve the power factor.

THD definition

THD is the total harmonic distortion present in a signal, defined as the ratio of the root-mean-square (RMS) amplitude of the total higher harmonic frequencies to the RMS amplitude of the fundamental frequency. Equation 1 expresses THD:

where Vn is the RMS value of the nth harmonic, and V1 is the RMS value of the fundamental component.

THD requirements have become stricter, especially in server applications, but meeting low THD requirements is more difficult than ever. The following methods can help reduce THD.

Make sure that the sensed signals are clean

To reduce THD, the first thing is to make sure that all of the sensed signals are clean. Because the sensed AC input voltage modulates the current reference, any spikes on the sensed AC signal will cause current reference distortion and affect THD. 

One common practice is to put a decoupling capacitor close to the analog-to-digital converter (ADC) pin of the controller and set the resistor-capacitor filter cutoff frequency about 10 times higher than the frequency you are interested in. If the sensed AC voltage is still noisy, you can use a software phase-locked loop (SPLL) [1] to generate an internal sine-wave signal in phase with the AC voltage, and then use that generated sine-wave signal to modulate the current reference. Since the SPLL-generated sine wave is clean, even if there is some noise on the sensed AC voltage, the current loop reference will still be clean.

For VOUT sensing, you can use a digital infinite impulse response filter, as shown in Equation 2, to process the sensed VOUT to further reduce noise; because the PFC voltage loop is slow, the extra delay caused by this digital filter is acceptable.

where k<1.

Oversampling

The PFC inductor current has switching ripples. The current-sensing circuit may not provide sufficient attenuation to this current ripple. If you sample this signal only once in each switching period, there is no perfect, fixed location where the signal represents the average current all of the time. To get a more accurate feedback signal, consider using an oversampling mechanism.

Figure 1 shows an example that evenly samples the current feedback signal eight times in every switching cycle, averages the results, and sends them to the control loop. This oversampling effectively averages the current ripple out such that the measured current signal gets closer to the average current value. Also, the controller becomes less sensitive to noise.

Figure 1 Oversampling eight times in every switching cycle to average the current ripple out in order to allow the measured current signal to get closer to the average current value. (Source: Texas Instruments)

Reduce the current spikes at AC zero crossing

Current spikes at AC zero crossing are an inherent issue for totem-pole bridgeless PFC. These spikes can be so big that it becomes impossible to pass THD specifications. Reference [2] analyzes the root cause of these spikes and provides a PWM soft-start algorithm to effectively reduce them, as shown in Figure 2.

Figure 2 PWM soft start after AC zero crossing to prevent current spikes common to totem-pole brideless PFCs.(Source: Texas Instruments)

In this algorithm, when VAC changes from a negative to a positive cycle after AC zero crossing, boost switch Q2 turns on first with a very small pulse width, then gradually increases to the duty cycle (D) generated by the control loop. A soft start on Q2 gradually discharges the switch-node drain-to-source voltage (VDS) to zero. Once Q2 soft start is complete, synchronous transistor Q1 starts to turn on. It begins with a tiny pulse width and gradually increases until the pulse width reaches 1–D. When Q2 soft start is complete and Q1 soft start begins, the low-frequency switch Q4 turns on.

The transition from the AC positive cycle to the negative cycle is similar. Turning off all of the switches at the end of each half AC cycle leaves a small dead zone at AC zero crossing. Figure 3 shows the test result.

Figure 3 Current waveforms without and with a PWM soft start: the traditional control method (a); PWM soft start (b). (Source: Texas Instruments)

Reduce voltage-loop effects

The PFC output voltage has double-line frequency ripples. Although the voltage loop compensator can reduce these ripples, it cannot totally remove them; there are still some ripples coupled to the current reference that then affect THD.

One way to reduce the effect of these ripples is to add a digital notch (band-stop) filter between the VOUT sensed signal and the voltage loop. This notch filter can effectively attenuate the double-line frequency ripple while still passing all other frequency signals, including the sudden VOUT change caused by the transient load. The load transient response will not be affected.

Another approach is to use VOUT at the AC zero-crossing value, or VOUT_ZC(t), as a voltage feedback signal; see Figure 4. Since VOUT_ZC(t) equals the average value of VOUT, and since it is a “constant” in steady state, using it as feedback signal can eliminate the double-line frequency ripple.

Figure 4 VOUT at the AC zero-crossing instant, using this method can eliminate the double-line frequency ripple. (Source: Texas Instruments)

To handle the load transient, use the voltage loop control law shown in Figure 5.

Figure 5 Using VOUT_ZC(t) as a feedback signal in the steady state. (Source: Texas Instruments)

If the instantaneous error is small, use the value at the AC zero-crossing instance, which is VOUT_ZC, and a small Kp, Ki for the voltage loop compensator Gv. When a load transient occurs, causing an instantaneous VOUT error greater than the threshold, use the instantaneous VOUT value and a large Kp, Ki for Gv to rapidly bring VOUT back to its nominal value.

Duty-ratio feedforward control

As the name suggests, duty-ratio feedforward control precalculates a duty ratio, then adds this duty ratio to the feedback controller. For a boost topology operating in continuous conduction mode, Equation 3 gives the duty ratio feedforward (dFF) as:

Figure 6 depicts the resulting control scheme. After using Equation 3 to calculate dFF, add dFF to the traditional average current-mode control output (dI). Then use the final duty ratio (d) to generate a PWM waveform to control PFC.

Figure 6 Average current-mode control with dFF. (Source: Texas Instruments)

Since dFF generates the majority of the duty cycle, the control loop only adjusts the calculated duty slightly. This technique can help improve THD for applications with a limited controller loop bandwidth.

Harmonic injection

In cases where a specific order of harmonics is too high, and the methods I’ve described still cannot meet the THD specification, a harmonic injection method [3] may resolve the problem. The basic idea of this method is to generate a sinusoidal signal with the same order of the harmonic that you want to compensate, and inject this signal into the PFC current control loop to compensate for that harmonic.

There are two ways to generate a sinusoidal signal. The first method is to use an SPLL to track the AC voltage and then generate the corresponding high-order harmonics. The second method is to generate a sine-wave table and then read the table element at a different speed to obtain different orders of sine waves [3]. Figure 7 shows a test result on a PFC that initially has high third- and fifth-order harmonics.

Figure 7 Harmonic injection to reduce third- and fifth-order harmonics. (Source: Texas Instruments)

Power factor definition

The power factor is the ratio of real power in watts to apparent power, which is the product of the RMS current and RMS voltage in volt amperes, as shown in Equation 4:

Ideally, the power factor should be 1; the load will then appear as a resistor to the AC source. In the real world, however, electrical loads not only cause distortions in AC current waveforms but also make the AC current either lead or lag with respect to the AC voltage, resulting in a poor power factor. For this reason, Equation 5 calculates the power factor by multiplying the distortion power factor by the displacement power factor:

where φ is the phase angle between the current and voltage, and THD is the total harmonic distortion of the current.

Equation 5 also shows that to improve the power factor, the first thing to do is to reduce THD. However, low THD does not necessarily mean that the power factor is high. If the PFC AC input current and AC input voltage are not in phase, even if the current is a perfect sine wave (low THD), φ will result in a power factor less than 1.

The phase difference between the input current and input voltage is mainly caused by the electromagnetic interference (EMI) filter used in the PFC. Figure 8 shows a typical PFC circuit diagram that consists of three major parts: an EMI filter, a diode bridge rectifier, and a boost converter.

Figure 8 Circuit diagram of a typical PFC comprising an EMI filter, a diode bridge rectifier, and a boost converter. (Source: Texas Instruments)

In Figure 8, C1, C2, C3 and C4 are EMI X-capacitors. Simplifying Figure 8 results in Figure 9, where C is now a combination of C1, C2, C3, and C4.

Figure 9 Simplified EMI filter combining the capacitances shown in Figure 8. (Source: Texas Instruments)

The X-capacitor causes the AC input current to lead the AC voltage, as shown in Figure 10. The PFC inductor current is , the input voltage is , and the X-capacitor reactive current is . The total PFC input current is , which is also the current from where the power factor is measured. Although the PFC current control loop forces to follow , the reactive current of leads by 90 degrees, which causes to lead . The result is a poor power factor.

This effect is amplified at a light load and high line, as takes more weight in the total current. As a result, it is difficult for the power factor to meet a rigorous specification.

Figure 10 X-capacitor causes the AC current to lead the AC voltage. (Source: Texas Instruments)

Fortunately, with a digital controller, you can resolve this problem with one of the following methods.

Delay the current reference

Since makes the total current lead the input voltage, you can force to lag  by some degree, as shown in Figure 11. The total current will then be in phase with the input voltage, improving the power factor.

Figure 11 Forcing to lag . (Source: Texas Instruments)

Since the current loop forces the inductor current to follow its reference, to let lag , the current reference needs to lag . To delay the current reference, a circulate buffer stores the measurement VAC results. Then, instead of using the newest input voltage VAC data, use previously stored VAC data to calculate the current reference for the present moment. The current reference will lag ; the current loop will then make  lag . This can compensate the leading X-capacitor  and improve the power factor.

The delay period needs dynamic adjustment based on the input voltage and output load. The lower the input voltage and the heavier the load, the shorter the delay needed. Otherwise will be over delayed, making the power factor worse than if there were no delay at all. To resolve this problem, use a look-up table to precisely and dynamically adjust the delay time based on the operating condition.

Subtract from the current reference

Since a poor power factor is caused mainly by the EMI X-capacitor , if you calculate  for a given X-capacitor value and input voltage and then subtract  from the total ideal input current to form a new current reference for the PFC current loop, you will get a better total input current that is in phase with the input voltage and can achieve a good power factor.

To explain in more detail, for a PFC with a unity power factor of 1, is in phase with . Equation 6 expresses the input voltage:

where VAC is the AC input peak value, and f is the AC frequency. The ideal input current then needs to be totally in phase with the input voltage, expressed by Equation 7:

where IAC is the input current peak value.

Equation 8 expresses the capacitor current:

Equation 9 comes from Figure 9:

Combining Equation 7, Equation 8, and Equation 9 results in Equation 10:

If you use Equation 10 as the current reference for the PFC current loop, you can fully compensate the EMI X-capacitor , achieving a unity power factor. In Figure 12, the blue curve is the waveform of the preferred input current, iAC(t), which is in phase with . The green curve is the capacitor current, iC(t), which leads  by 90 degrees. The red curve is iAC(t) ‒ iC(t). In theory, if the PFC current loop uses this red curve as its reference, you can fully compensate the EMI X-capacitor  and increase the power factor.

Figure 12 New current reference. (Source: Texas Instruments)

Equation 10 requires a cosine waveform cos (2πƒt). To get this cosine waveform, use an SPLL to generate an internal sine wave synchronized with the input voltage. For microcontrollers that cannot perform trigonometric calculations, reference [4] describes another way to calculate iC(t).

Reduce THD and improve PF

If you need to reduce THD and improve the power factor, choose one or a combination of the methods discussed here. In the next installment of this series, I will talk about how to improve efficiency, limit re-rush current, implement e-metering, and reduce PFC bulk cap with a baby boost converter.

Related Content

• How to design a digital-controlled PFC, Part 1
• How to design a digital-controlled PFC, Part 2
• Power Tips #116: How to reduce THD of a PFC
• Power Tips #124: How to improve the power factor of a PFC

References

1. Bhardwaj, Manish. “Software Phase Locked Loop Design Using C2000 Microcontrollers for Single Phase Grid Connected Inverter.” Texas Instruments application report, literature No. SPRABT3A, July 2017.
2. Sun, Bosheng. “How to Reduce Current Spikes at AC Zero Crossing for Totem-Pole PFC.” Texas Instruments Analog Design Journal article, literature No. SLYT650, 4Q 2015.
3. Sun, Bosheng. “A Harmonic Injection Method to Reduce Harmonics and THD for PFC.” Power Electronics News, Nov. 20, 2023.
4. Sun, Bosheng. “Increase power factor by digitally compensating for PFC EMI-capacitor reactive current.” Texas Instruments Analog Design Journal article, literature No. SLYT673, 2Q 2016.

The post How to design a digital-controlled PFC, Part 3 appeared first on EDN.

Sharing this as a heads-up for anyone considering JLCPCB's assembly service. JLCPCB lost parts I pre-purchased through their own platform, produced boards with cold solder defects, then shipped the defective incomplete boards two days after I explicitly told them not to ship. Three weeks later I still have no working product. Their support has been like talking to a bot. I've been asked three times to arrange a local repair despite explaining each time that it's not possible — they never populated an SMD component that they lost, and you can't fix that with a soldering iron. Each response only acknowledges one issue and ignores the rest. When I asked for a replacement order, I was told it "goes beyond their normal compensation policy" because of their internal material costs and production backlogs. Every reply is vague — they "may" arrange a return, they "may" apply for a coupon. No commitments, no timeline, nothing concrete. I'm also now sitting with £81 in import charges on a defective package I never asked to receive, currently stuck in a courier warehouse because nobody knows what to do with it. Their bare PCB service is fine. But if you're relying on their assembly service for anything with a real deadline, understand that when they make a mistake, their process is designed to exhaust you into accepting it rather than actually fixing it. submitted by /u/gogosomewhere [link] [comments]

After years of having my multimeters die exactly when I most need then, I finally made good use of a vape battery, a TP4056 charger, a 9v boost and a female usb-c on a cheap red multimeter. Not ready to do on my good meter, but I am very happy with this little mod. submitted by /u/nomoreimfull [link] [comments]

Efficient Power Conversion Corp (EPC) of El Segundo, CA, USA — which makes enhancement-mode gallium nitride on silicon (eGaN) power field-effect transistors (FETs) and integrated circuits for power management applications — has launched a new generation of 100V integrated GaN power-stage ICs (EPC23108, EPC23109, EPC23110 and EPC23111) targeting high-performance motion and power systems such as humanoid robots, drones, and other compact battery-powered platforms. The devices are designed to simplify implementation and improve operational robustness in real-world environments while preserving the efficiency and power density advantages typical of integrated GaN technology...

💡 Вебінар «Актуалізація профілів авторів у цифровому середовищі» kpi пт, 04/03/2026 - 12:32

Візит експрем’єр-міністра Республіки Болгарія Кирила Петкова kpi пт, 04/03/2026 - 11:48

Краса та ніжність на виставці у ЦКМ Інформація КП пт, 04/03/2026 - 10:00

submitted by /u/AdiDevMC [link] [comments]

Hello everyone, this is the first version of my function generator. I'm looking for recommendations! Before you comment: - I made it out of discrete parts because the goal was learning more than immediate results. - I'm a second year ECE so many mistakes will be expected. I'm still in Electronics I and learning about DC/low frequency circuits. - I plan to use 50Ohm input impendance but need a beefier power supply and maybe transistors. (currently using 2 9V rechargable batteris for sine and 1 for square) - I only have that oscilloscope - I'll only use it - Used a Pi Pico W in order to add in the future more functions. - Code was ai generated with my tweaks and fixes on it. As much as it hurts to say it's the truth as I preferred to work on hardware for now. I do know C++ and will learn it better. - KiCad files don't include the square circuit as it's not yet perfect*. Project Goals (v1.0): - Arbitrary wave generation (left it behind for now as it's just another R-2R - Sinewave and squarewave generation up to 1MHz. - 1k Ohm input impendance * Sadly I don't have a square wave photo (and won't be home for 2 weeks) but it was perfect up to 200kHz. After that the duty cycle got smaller but in terms of noise/rounding it was pretty good. Plus the rise time at 1MHz wasn't perfect but pretty okay. If anyone has any ideas lmk. Way it works: - Sine: R-2R -> active filter -> 3RC LPF and one RC HPF for dc cutoff -> Amp (+9V, -9V) -> Buffer - Square: PWM on/off -> amp & buffer (9V, 0V) Images: Sinewave physical circuit Sinewave output Squarewave circuit (input is 3.3v square pwm) Sinewave schematic For way more info: GitHub repo Edit: Not sure why Vpp is 120V pretty sure had x1 on the oscilloscope or something. submitted by /u/S4vDs [link] [comments]

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