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Peltz oscillator with injection locking

Oscillator injection locking is an interesting subject; however, it seems to be a forgotten circuit concept that can be beneficial in some applications.

Wow the engineering world with your unique design: Design Ideas Submission Guide

This design idea shows an application of the built-in Bode capability within many modern low-cost DSOs such as the Siglent SDS814X HD using the Peltz oscillator as a candidate for investigating injection locking [1], [2], [3].

Figure 1 illustrates the instrument setup and device under test (DUT) oscillator schematic with Q1 and Q2 as 2N3904s, L ~ 470 µH, C ~ 10 nF, Rb = 10K, Ri = 100K and Vbias = -1 VDC. This arrangement and component values produce a free running oscillator frequency of ~75.5 kHz

Figure 1 Mike Wyatt’s notes on producing a Peltz oscillator and injector locking setup where the arrangement and component values produce a free running oscillator frequency of ~75.5 kHz.

Analysis and measurements

As shown in Figure 2, the analysis from Razavi [2] shows the injection locking range (± Δfo) around the free running oscillator frequency fo. Note the locking range is proportional to the injected current Ii. The component values shown reflect actual measurements from an LCR meter.

Figure 2 Mike Wyatt’s notes on the injection-locked Peltz oscillator showing the injection locking range around the free running oscillator frequency fo.

This analysis predicts a total injecting locking range of 2*Δfo, or 2.7 kHz, which agrees well with the measured response as shown in Figure 3.

Figure 3 The measured response of the circuit shown in Figure 1 showing an injection locking range of roughly 2.7 kHz.

Increasing the injection signal increases the locking range to 3.7 kHz as predicted, and measurement shows 3.6 kHz as shown in the second plot in Figure 4.

Figure 4 The measured response of the circuit shown in Figure 1 where increasing the injection signal increases the locking range to 3.7 kHz.

Note the measured results show a phase reversal as compared to the illustration notes (Figure 2) and the Razavi [2] article. This was due to the author not defining the initial phase setup (180o reversed) in agreement with the article and completing the measurements before realizing such!!

Injection locking use case

Injection locking is an interesting subject with some uses even in today’s modern circuitry. For example, I recall an inexpensive arbitrary waveform generator (AWG) which had a relatively large frequency error due to the cheap internal crystal oscillator utilized and wanted the ability to use a 10 MHz GPS-disciplined signal source to improve the AWG waveform frequency accuracy. Instead of having to reconfigure the internal oscillator and butcher up the PCB, a simple series RC from a repurposed rear AWG BNC connector to the right circuit location solved the problem without a single cut to the PCB! The AWG would operate normally with the internal crystal oscillator reference unless an external reference signal was applied, then the oscillator would injection lock to the external reference. This was automatic without need for a switch or setting a firmware parameter, simple “old school” technique solving a present-day problem!

 Michael A Wyatt is a life member with IEEE and has continued to enjoy electronics ever since his childhood. Mike has a long career spanning Honeywell, Northrop Grumman, Insyte/ITT/Exelis/Harris, ViaSat and retiring (semi) with Wyatt Labs. During his career he accumulated 32 US Patents and in the past published a few EDN Articles including Best Idea of the Year in 1989.

References

1. “EEVblog Electronics Community Forum.” Injection Locked Peltz Oscillator with Bode Analysis, www.eevblog.com/forum/projects/injection-locked-peltz-oscillator-with-bode-analysis. 
2. B. Razavi, “A study of injection locking and pulling in oscillators,” in IEEE Journal of Solid-State Circuits, vol. 39, no. 9, pp. 1415-1424, Sept. 2004, doi: 10.1109/JSSC.2004.831608. 
3. Wyatt, Mike. “Simple 5-Component Oscillator Works below 0.8V.” EDN, 3 Feb. 2025, www.edn.com/simple-5-component-oscillator-works-below-0-8v/.

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The post Investigating injection locking with DSO Bode function appeared first on EDN.

While finetuning its products and manufacturing process roadmap, Intel has realized that there are no quick fixes. After a briefing from Intel co-CEOs Michelle Holthaus and David Zinsner on upcoming CPUs and a slowdown in the ramp of the 18A node, Alan Patterson caught up with industry analysts to take a closer look at Intel’s predicament. He spoke with them about delayed CPU launches, the lack of an AI story, and the fate of Intel Foundry.

Read the full story at EDN’s sister publication, EE Times.

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The post Intel comes down to earth after CPUs and foundry business review appeared first on EDN.

Battery-powered applications, which have become indispensable over the last decade, require a certain level of protection to ensure safe use. This safety is provided by the battery management system (BMS). The BMS monitors the battery and possible fault conditions, prevents any hazardous situation due to the battery or its surroundings, and ensures that there is an accurate estimation of the battery’s remaining capacity or the level of battery degradation.

The main structure of a BMS for a low- or medium-voltage battery is commonly made up of three ICs, as described below:

1. Battery monitor and protector: Also known as the analog front-end (AFE), the battery monitor and protector provides the first level of protection since it’s responsible for measuring the battery’s voltages, currents, and temperatures.
2. Microcontroller unit (MCU): The MCU, which processes the data coming from the battery monitor and protector, commonly incorporates a second level of protection, including monitoring thresholds.
3. Fuel gauge (FG): The fuel gauge is a separate IC that provides the state-of-charge (SOC), state-of-health (SOH) information and remaining runtime estimates, as well as other user-related battery parameters.

Figure 1 The BMS architecture displays the key three building blocks. Source: Monolithic Power Systems

Figure 1 shows the main structure of a complete BMS for low- or medium-voltage batteries. The fuel gauge can be a standalone IC, or it can be embedded in the MCU. The MCU is the central element of the BMS, taking information from both the AFE and fuel gauge and interfacing with the rest of the system.

While three main components constitute the BMS, using these components without any additional consideration is not enough to ensure that the system meets the safety level required by certain industries. This article will explain the role that functional safety plays in non-automotive battery management systems and how to achieve the required safety level.

Functional safety introduction

Functional safety is a branch of overall safety focused on reducing the risk produced by hazardous events due to a functional failure of an electric/electronic (E/E) system. The goal is to ensure that the residual risk is within an acceptable range.

In recent years, the increasing use of E/E systems in different fields such as automotive, machinery, medicine, industry, and aviation has been accompanied by a greater emphasis on functional safety. These changes have led to the development of different functional safety standards.

ISO 13849, titled “Safety of machinery – Safety related part of control systems”, is a functional safety standard focused on the safety-related parts of control systems (SRP/CS) in the machinery field. This is a field that includes a wide spectrum of applications, from generic industrial machinery to mopeds and e-bikes. ISO 13849 defines different safety levels as performance level (PL), which range from PLa (lower safety level) to PLe (higher safety level).

This safety standard defines an accurate process for risk evaluation and reduction. It proposes a simplified method to determine the achieved PL based on three parameters: category, mean time to dangerous failure (MTTFD), and average diagnostic coverage (DCAVG), which is calculated by averaging all the DC associated to the different safety measures applied in the system.

The category is a classification of an SRP/CS that describes its resistance to faults and the subsequent behavior in the event of a fault condition. There are 5 categories (B, 1, 2, 3, and 4).

Architecture has the biggest impact on the category. The basic architecture of an SRP/CS is composed of three functional blocks: an input, a logic block, and an output (Figure 2). Figure 2 corresponds with the architecture proposed for category B and category 1, and it’s called a “single-channel” architecture. A single-channel architecture is considered the most basic architecture to implement the nominal functionality of the SRP/CS, but it’s not intended for any diagnostic functionality.

Figure 2 The above architecture is proposed for category B and category 1. Source: Monolithic Power Systems

Category B and 1 rely on the reliability of their components (MTTFD) to ensure the integrity of the safety functions. If a component implementing the safety function has a failure, a safe state can no longer be guaranteed, as no diagnostics are implemented (DCAVG = 0).

For category 2, the proposed architecture is called “single-channel tested.” The base of this architecture is the same as the single-channel architecture, but with an added test equipment block that can diagnose whether the functional channel is working correctly. If a component implementing the safety function has a failure, the safety function is not carried out; however, a safe state can be achieved if the failure is diagnosed by the test equipment.

For category 3 and category 4, the proposed architecture is called “redundant channels,” which is implemented with two independent functional channels that can diagnose issues on the other channel. If a component implementing the safety function has a failure, the safety function can still be carried out by the other channel. Designers should select the SRP/CS category based on the targeted safety level of each safety function.

Achieving functional safety step-by-step

The ISO 13849 standard defines an iterative process during which the SRP/CS design is evaluated to determine the achieved PL and check whether that safety level is sufficient or must be improved in a new loop. The process includes three different methods for risk reduction: risk reduction via safe designs measures, risk reduction via safeguarding, and risk reduction via information for use. ISO 13849 supports risk reduction via safeguarding (Figure 3).

Figure 3 ISO 13849 supports risk reduction via safeguarding. Source: Monolithic Power Systems

The safeguarding process starts by defining the safety functions of the SRP/CS, in which the required performance level (PLr) is defined after the risk analysis is conducted. The PLr is the target PL of the SRP/CS for each safety function.

The next step includes designing the SRP/CS for the specified safety requirements. This entails considering the possible architecture, the safety measures to implement, and finalizing the design of the SRP/CS to perform the relevant safety functions.

Once the SRP/CS is designed, evaluate the achieved performance level for each safety function. This is the core step of the entire safeguarding process. To evaluate the achieved PL, define the category and then calculate the MTTFD and DCAVG of the SRP/CS for each individual safety function.

The MTTFD is calculated per channel, and it has three levels (Table 1).

Table 1 MTTFD, calculated per channel, has three levels. Source: Monolithic Power Systems

Table 2 shows the four levels for defining the DC of each diagnostic measure.

Table 2 There are four levels for defining the DC of each diagnostic measure. Source: Monolithic Power Systems

The achievable PL can be determined using the relevant parameters (Table 3).

Table 3 Relevant parameters help determine the achievable PL. Source: Monolithic Power Systems

The achievable PL can only be confirmed when the remaining requirements and analyses defined by the standard are implemented in the design. These requirements must comply with systematic failures management, common cause failure (CCF) analysis, safety principles and software development, if applicable.

Once this process is complete, the PL achieved by the SRP/CS for a concrete safety function should be verified against the PLr. If PL < PLr, then the SRP/CS should be redesigned, and the PL evaluation process must begin again. If PL ≥ PLr, then the SRP/CS has achieved the required safety level, and validation must be executed to ensure the correct behavior through testing. If there is an unexpected behavior, the SRP/CS should be redesigned. This process should be reiterated for each safety function.

Functional safety level according to each market

Battery-powered devices are used in countless markets, and each market demands different functional safety specifications according to how dangerous a failure could be for humans and/or the environment. Table 4 shows the functional safety level required by some of the main markets. Note that these levels are constantly changing and may be different depending on each engineering team’s design.

Table 4 This is how PL is determined based on market. Source: Monolithic Power Systems

Although these are the current performance level market expectations, electromobility and certain energy storage applications may move into PLd due to the constant issues in battery-powered devices around the world. For example, faulty energy storge applications have resulted in fires in U.S. energy storage system (ESS) facilities. In U.K., more than 190 persons have been injured, and eight persons have been killed by fires sparked by faulty e-bikes and e-scooters.

All these events could have been prevented by a more robust and reliable system. The constant need for increasing safety levels means it is vital to have a scalable solution that can be implemented across different performance levels.

A functional safety design proposal

Take the case of an ISO 13849-based BMS concept that Monolithic Power Systems (MPS) has developed by combining an MCU with its MP279x family of battery monitors and protectors. This system is oriented to achieve up to PLc safety level for a certain set of safety functions (SFs), as shown in Table 5. PLr determination is dependent on the risk analysis, in which small variations can take place, as well as the application in which the BMS is used.

Table 5 See the defined safety functions for the BMS concept. Source: Monolithic Power Systems

The solution proposed by MPS to achieve PLc can meet category 2 or category 3—depending on each safety function—as for certain safety functions. There is only a single input block and for others, there are redundant input blocks.

Figure 4 shows how to implement SF2 and SF4 to prevent the battery pack from over-charging and under-charging. In the implementation of the SRP/CS, there are two logic blocks: the battery monitor and protector (logic 1) and the MCU (logic 2). These logic blocks are used to diagnose correct functionality of different parts in the design.

Figure 4 Here is how to implement SF2 and SF4. Source: Monolithic Power Systems

The implementation of single or duplicated input is determined by the complexity and cost in each case. To ensure that the safety functions for a single input are compliant with PLc, additional safety measures can be taken to increase the diagnostic capability; an example is a cell voltage plausibility check to verify that the cell voltage measurements are correct.

Functional safety used to be relevant for automotive products, but nowadays most modern markets demand the manufacturer to comply with a functional safety standard. The best-known safety standard for non-automotive markets is ISO 13849, a system-level standard that ensures an application’s safety and robustness.

Miguel Angel Sanchez is applications engineer at Monolithic Power Systems.

Diego Quintana is functional safety engineer at Monolithic Power Systems.

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The post Functional safety in non-automotive BMS designs appeared first on EDN.

Remember my April 2023 teardown of Spotify’s now-defunct Car Thing?

Ditch the touchscreen LCD, broaden functionality and that’s Amazon’s Echo Auto in a nutshell:

Shown here and introduced in mid-2019 is the first-generation version of the product, which I’ll be tearing down today. It originally sold for $49.99 but was initially promo-priced at half that amount ($24.99), which is how it came to be in my possession that same summer. The second-gen successor, introduced three years (and three months) later with shipments beginning in mid-December 2022, was smaller, with beefier mounting options, equivalent claimed input-sound quality (in spite of fewer integrated mics) and a supposed superior sonic output, along with a permanent 24.99 price cut. It’s still available for purchase:

Considering that the first-gen Echo Auto has been sitting on my shelf for more than 5 years now awaiting my dissection attention, the beat-up condition of its packaging, as-usual accompanied by a 0.75″ (19.1 mm) diameter U.S. penny for size comparison purposes, would be understandable…except that it’s looked like this since it first showed up at my front door!

Rip off the retaining tape and flip open the top flap:

and the contents come into view.

Post-removal, here’s our patient, alongside the similarly clear plastic-clad (at least for the moment) dashboard mount:

the “cigarette lighter” 12V socket-based power supply, flanked by (on the left) a 3.5mm TRS extension cable and (on the right) the USB-A to micro-USB power cable, all three of which I’ll hold on to for future reuse:

and, of course, a few slivers of documentation:

Next, a couple of additional looks at the adhesive dash mount (and its accompanying preparatory dashboard-cleaning handi wipe), now free of its clear plastic sarcophagus:

and the power adapter, with a handy included second USB-A jack, and decent aggregate output:

With the contents removed and its insides now ostensibly empty, the box still seems hefty, but I confirmed that there was nothing left within. Must be all those folded cardboard layers:

And now for some initial perspectives on our patient, with dimensions of 3.3” x 1.9” x 0.5” (85 mm x 47 mm x 13.28 mm) and a weight of 1.6 oz (45 grams). Front:

The left “mute” button, by the way, turns red when active, as with other Echo devices, as does the more general multicolor device-status light bar along the bottom edge:

The device top is comparatively bland, although there is that inside access-tempting seam:

The sides are more interesting. Along the right are the 3.5mm auxiliary analog audio output and the micro-USB power connector. The former was a key motivation for me to initially buy the Echo Auto, as none of my vehicles have integrated Bluetooth, far from Apple’s CarPlay or Google’s Android Auto services—only my wife’s newer car does—but their sound systems all have AUX inputs.

And on the left? No, that’s not a SD card slot. Believe it or not, it’s the aperture for the integrated speaker, pointing toward the vehicle’s driver (at least sometimes):

Finally, the device backside, revealing (among other things) the FCC ID (2ALV8-4833) and magnetic dash mount inset (I trust there’s metal inside, on the other side of the chassis):

Speaking of “inside”, let’s get to it. A preparatory peek underneath one of the rubber feet seemingly wasn’t promising:

So, I turned my attention to the aforementioned top side seam. The first “spudger” I tried slipped inside fairly easily but was too flimsy to make any separation headway:

Its beefier Jimmy sibling, however, was no more successful:

On a hunch, I revisited those feet. That grey piece of plastic you saw underneath the one in the earlier photo? Turns out, it pops out too:

And underneath each of the plastic pieces is a hex screw head begging for attention:

That’s more like it:

FWIW, as it turns out from my subsequent research, I wasn’t the only one initially flummoxed!

There’s that piece of metal I’d previously forecasted would be on the other side of the dashboard mount inset. Below it, along the bottom edge, is a portion of the light guide assembly (presumably associated with a to-be-seen row of LEDs on the PCB):

And here’s our first glimpse of the system’s guts:

On the left (right when viewed from the front; remember that we’ve so far removed the back panel) is the micro-USB power input, with the 3.5 mm audio jack above it. Along the bottom are—I told you so—a row of 11 multicolor LEDs. At the top is the PCB-embedded Bluetooth antenna. And on the right? That, believe it or not, is the mono speaker! Let’s get it outta there:

Lest there be any doubt as to its magnet-inclusive acoustic identity:

And now for some closeups, with perspectives oriented per the transducer as originally installed in the previous photo. Right side, where the sound comes out; I seriously doubt it “goes to 11”:

Front:

Left side:

Back, exposing the speaker’s electrical contacts:

And finally, the top:

and bottom:

With the speaker removed, you can now see the PCB-resident “spring” contacts that mate up with those on the speaker. Note, too, that the PCB holes corresponding to mounting pins on the speaker backside are foam-reinforced, presumably to suppress vibration while in operation:

And now let’s get the PCB out of there, a thankfully easier process than what’d previously been necessary to get our first glimpse of it, as it now lifts right out of the remaining chassis half:

The stuck-on RFID tag inside the front chassis half is an interesting story in and of itself. As this blogger also postulates (in addition to identifying the source—Inpinj—of the IC connected to the comparatively massive antenna), I believe that it finds use in uniquely associating the device with your Amazon account prior to its shipment to you. To wit, I happened to notice, in reviewing my Amazon order history to refresh my memory of when I bought the Echo Auto and what I paid for it, that the device serial number was also included in the relevant transaction listing. And at the bottom is the other portion of the light guide assembly:

Here’s the already-seen PCB backside, now free of its previous plastic chassis surroundings:

And here’s the first-time glimpsed PCB front side:

Let’s first get rid of that rubber gasket, which thankfully peeled off easily:

Note the LEDs straddling the left-side switch, which generate the red “mute” indication. Note, too, eight total circular apertures for the microphone array, one in each corner of each of the two switches. And as for the ICs between the switches, let’s zoom in:

Unfortunately, I had no luck in identifying any of these; I’m once again hopeful that insightful readers can fill in the missing pieces. The one at the bottom (U10), when correctly oriented (it’s upside-down marked in the photo) has what looks to be an “OXZ” company logo stamped in the upper left corner. The three-line product marking next to it looks like this:

L16A
0225
ZSD838A

I found similar markings (albeit with second-line deviations) on an IC inside a 2018-2019 13” Apple MacBook Air, within a Facebook post which I stumbled across thanks to Google Image Search, but that’s all I’ve got. Above it are two ICs (U2 and U6) identically marked as follows:

YE08
89T

which may be 8-bit bidirectional voltage-level translators, specifically Texas Instruments’ TXB0108. And in U10’s upper right corner is another (U9) with the following two-line marking:

T3182
3236A

Again…

Let’s flip the PCB back over to its backside and see if we have any better luck. Step one is to get those two Faraday Cages’ tops off:

That’s better:

The IC at far left (U20), next to a wire-wound inductor whose guts seem to have been inadvertently exposed by the spudger while removing the cage, is labeled thusly (and faintly so):

25940A
TI 89I
AE24

“TI” stands for “Texas Instruments”, I’m pretty confident, reflective of the longstanding partnership between that supplier and Amazon also noted in several of my past Echo product dissections. And Texas Instruments does have a “25940” in its product line, specifically the TPS25940, the “eFuse Power Switch”, a “compact, feature-rich power management device with a full suite of protection functions, including low power DevSleep support”. If that’s actually what this chip is, its proximity to the micro-USB power input therefore makes sense. But the product page also claims that the TPS25940 is intended for use in SSDs. Hmm…

Above and to the right of it is another chip with “TI” in the markings (U14), but the first line thankfully makes its function more obvious, at least as far as I’m guessing:

DAC
3203I
TI 88J
PL49

This, I believe, is Texas Instruments’ TLV320DAC3203 “stereo” audio DAC with a stereo 125-mW headphone driver and audio processing. Proximity is again part of the probable identity tip-off here, since it’s near the analog audio output. Plus, of course, there’s the first-line “DAC” mark…

Move further to the right and the next large(r) IC you encounter (U19), also seemingly chipped in one corner during my clumsy cages-removal surgery, has the following two-line primary markings (along with, above them, a combo mysterious swirl followed by a seeming QR code):

W902B108
SR3F2

Google searches on the markings proved fruitless but, based on some other research I’ve done on this system, I’m still going to take a guess. The Amazon product page indicates that in addition to the main system SoC (hold that thought), there’s also an “Intel Dual DSP with Inference Engine” inside. The relevant DeviWiki product page further clarifies that it’s an “Intel Quark S1000 Processor.” Indulge me in a brief history diversion: a bit more than a decade ago, Intel announced its Quark line of defeatured 32-bit x86 processors (even more so than its Atom CPUs) for wearables and other cost- and power-sensitive applications. The Quark family, which Intel obsoleted in 2019, also included at least one coprocessor, the S1000, which embedded two Cadence Tensilica LX6 DSP cores. Intended for speech recognition, I assume that the S1000 also handled echo cancellation, background noise suppression and other array mic functions in this particular design. And I’m also guessing that, although there’s no Intel logo mark, it’s this chip.

Now for the main system SoC (U23), which is to the right of the previous “mystery chip” and is thankfully more easily identifiable. It’s MediaTek’s MT7697, introduced in 2016 and described as a “highly integrated 1T1R 2.4GHz Wi-Fi/Bluetooth 4.2 application processor with an Arm Cortex-M4 and a power management unit”, MediaTek being another supplier with a longstanding Amazon relationship.

Which leads us to the last chip I’ll showcase, to its right, with a two-IC PCB identifier (U17/U18). At first, I thought the “MT” mark might also indicate MediaTek sourcing but, given that the MT7697 already also handles Bluetooth and power management functions, I couldn’t think of anything else this one could tackle. But then I remembered I hadn’t yet mentioned memory, either volatile or nonvolatile. This insight led me to suspect that “MT” probably instead stands for “Micron Technology” and that this is a stacked module containing both DRAM and flash memory (capacities and specific technology types and generations unknown).

In closing, I’ll (re)point out two other aspects of this side of the PCB; the eight MEMS microphones whose apertures you saw earlier on the other side, and the PCB-embedded top-edge Bluetooth antenna that I first noted when the PCB was still chassis-bound. And with that, having just passed through 2,000 words, I’ll wrap up with a reiteration of the invitation to assist me with any/all of the ICs I was unable to ID, and/or to share any other insights or other thoughts, in the comments. Thanks as always in advance!

—Brian Dipert is the Editor-in-Chief of the Edge AI and Vision Alliance, and a Senior Analyst at BDTI and Editor-in-Chief of InsideDSP, the company’s online newsletter.

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The post Amazon’s Echo Auto Assistant: Legacy vehicle retrofit-relevant appeared first on EDN.

Vietnam has scored another victory in its bid to move beyond back-end assembly and packaging and establish an IC design and testing presence. Mixel, an analog and mixed-signal IP developer, is opening a design shop in Da Nang, Vietnam.

The San Jose, California-based design house provides interface IP solutions for MIPI, LVDS, and other multi-standard SerDes. It was the first IP provider to demonstrate silicon for MIPI D-PHY, MIPI C-PHY, and MIPI M-PHY. Mixel’s new design office in Vietnam—the first in Asia—will contribute to its IP development work.

Figure 1 A design house serving high-speed mixed-signal IP market will help develop engineering talent in Vietnam. Source: Mixel

It follows Vietnam’s inking of strategic pacts with two large EDA houses—Cadence and Synopsys—to advance design talent and cultivate a culture of semiconductor startups. Vietnam National Innovation Center (NIC), currently setting up the infrastructure for an IC design incubation center at Hoa Lac High-Tech Park in Hanoi, has joined hands with Cadence to accelerate IC design activities.

As part of this program, Cadence provides access to its design tools to academic institutes selected by NIC. University students and professors can use the Cadence tools and online training suites to gain real-world IC design expertise. Cadence is also introducing internships and job opportunities to Vietnamese engineers who are trained at NIC.

Next, Synopsys provides training licenses and educational resources to help NIC set up the chip design incubation center. Here, NIC investes in prototyping and emulation infrastructure to cultivate the IC design workforce in collaboration with Synopsys. The Sunnyvale, California-based EDA house also provides prototyping and emulation tools for software and hardware co-design in system-on-chip (SoC) devices.

Figure 2 Chip designers at NIC’s incubation center will be trained on the latest IC design tools. Source: Synopsys

Vietnam is striving to seize the moment amid trade tensions between China and the United States. In the so-called “China+1” investment strategy, Vietnam is emerging as a major beneficiary, so it wants to complement IC design with the existing back-end manufacturing, testing, and packaging businesses.

If Vietnam is successful in its bid to develop a vibrant IC design industry, it will also integrate the country into the global semiconductor ecosystem.

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The post Vietnam’s pivot to the IC design world appeared first on EDN.

A low-noise chip-scale atomic clock (LN-CSAC), the SA65-LN from Microchip, features a profile height of less than 0.5 in. (12.7 mm). Aimed at aerospace and defense applications where size, weight, and power are critical, the SA65-LN delivers precise and stable timing, along with low phase noise and atomic clock stability.

Based on Microchip’s Evacuated Miniature Crystal Oscillator (EMXO) and integrated into a CSAC, the SA65-LN consumes under 295 mW. It also operates within a temperature range of -40°C to +80°C, maintaining its frequency and phase stability. Low power consumption and a wide temperature range enable battery-powered operation under extreme conditions.

The LN-CSAC combines the stability of an atomic clock with the precision of a crystal oscillator in a compact design. The EMXO offers low phase noise of <−120 dBc/Hz at 10 Hz and an Allan Deviation (ADEV) of <1E-11 at a 1-second averaging time. The atomic clock provides ±0.5 ppb initial accuracy, frequency drift of <0.9 ppb/month, and temperature-induced errors of <±0.3 ppb.

The SA65-LN is available now in production quantities. It is supported by Microchip’s Clockstudio software tool, a GUI, and developer kit.

SA65-LN product page

Microchip Technology

Find more datasheets on products like this one at Datasheets.com, searchable by category, part #, description, manufacturer, and more.

The post Chip-scale atomic clock shrinks profile height appeared first on EDN.

Infineon has added five isolated gate drivers to its EiceDriver family optimized for driving IGBTs and SiC MOSFETs. AEC-qualified and ISO 26262-compliant, these third-generation drivers are well-suited for traction inverters in both cost-effective and high-performance xEV platforms. Additionally, they support Infineon’s HybridPack Drive G2 Fusion module, a plug-and-play power module that combines the company’s Si and SiC technologies.

The 1EDI302xAS series supports IGBTs up to 1200 V, while the 1EDI303xAS series is suited for SiC MOSFETs up to 1200 V. With an output stage of 20 A, the 1EDI3025AS, 1EDI3026AS, and 1EDI3035AS can drive inverters of all power classes up to 300 kW. The 1EDI3028AS and 1EDI3038AS variants have an output stage of 15 A, useful for entry-level battery EV and plug-in hybrid EV inverters. The gate drivers provide reinforced insulation per VDE 0884-17:2011-10, ensuring safe isolation.

All of the single-channel drivers are equipped with a configurable soft turn-off feature for enhanced short-circuit performance. Monitoring functions include overcurrent protection and an integrated self-test for desaturation protection. A continuously sampling 12-bit delta-sigma ADC with an integrated current source can read the voltage directly from temperature measurement diodes or an NTC.

Samples of the 1EDI3025AS, 1EDI3026AS, 1EDI3028AS, 1EDI3035AS, and 1EDI3038AS isolated gate drivers are available now.

EiceDriver product page

Infineon Technologies 

Find more datasheets on products like this one at Datasheets.com, searchable by category, part #, description, manufacturer, and more.

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Keysight provides an end-to-end LPDDR6 memory design and test platform that improves device and system validation. It includes new test automation tools necessary for advancing AI, especially in mobile and edge applications.

Based on the UXR oscilloscope and M8040A bit error ratio tester, the complete setup includes transmitter and receiver test apps paired with the Advanced Design System (ADS) Memory Designer and EDA software. The LPDDR6 memory standard’s combination of high performance and power efficiency makes it particularly suitable for AI and machine learning workloads, high-speed digital computing, automotive systems, and data centers.

When used for transmitter testing, the platform reduces validation time with fully automated compliance testing and characterization. Engineers can analyze device BER performance with extrapolated eye mask margin testing and achieve accurate signal measurements directly from BGA packages with specialized de-embedding capabilities.

For receiver testing, the setup validates designs using with BER test methodology and pinpoints performance issues by testing against multiple jitter, crosstalk, and noise scenarios. It also ensures interoperability with both device and host controller validation.

The receiver and transmitter solution made its public debut at DesignCon 2025.

Keysight Technologies

Find more datasheets on products like this one at Datasheets.com, searchable by category, part #, description, manufacturer, and more.

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The NEX30606 step-down converter from Nexperia delivers up to 600 mA of output current with an operating quiescent current of just 220 nA. Supporting input voltages from 1.8 V to 5.0 V, the converter offers 16 resistor-settable fixed output voltages and uses constant on-time control for fast transient response.

Ultra-low quiescent current makes the NEX30606 well-suited for consumer wearables like hearing aids, medical sensors, patches, and monitors. It can also be used in battery-powered industrial applications, including smart meters and asset trackers. The converter provides greater than 90% switching efficiency for load currents ranging from 1 mA to 400 mA. Additionally, it has only 10 mV of output voltage ripple when stepping down from 3.6 VIN to 1.8 VOUT.

Nexperia also offers the NEX40400, a step-down converter that combines high efficiency with an operating quiescent current of 60 µA typical. It provides up to 600 mA of output current from a wide 4.5-V to 40-V input voltage range. The device employs pulse frequency modulation for high efficiency at low to mid loads and spread spectrum technology to minimize EMI. Target applications include industrial distributed power systems and grid infrastructure.

Visit the NEX30606 and NEX40400 product pages to check pricing and availability.

NEX30606 product page 

NEX40400 product page 

Nexperia

Find more datasheets on products like this one at Datasheets.com, searchable by category, part #, description, manufacturer, and more.

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Wolfspeed introduced its Gen 4 SiC MOSFET platform, supporting long-term roadmaps for high-power, application-optimized products. Gen 4 offerings include power modules, discrete components, and bare die available in 750-V, 1200-V, and 2300-V classes.

According to Wolfspeed, it is the only producer with both silicon carbide material and silicon carbide device fabrication facilities based in the U.S. This factor is becoming increasingly important under the new U.S. Administration’s increased focus on national security and investment in U.S. semiconductor production.

The Gen 4 platform was designed to improve system efficiency and prolong application life, even in the harshest environments. It is expected to deliver performance enhancements in high-power automotive, industrial, and renewable energy systems, with key benefits including: 

• Holistic system efficiency: Delivering up to a 21% reduction in on-resistance at operating temperatures with up to 15% lower switching losses.
• Durability: Ensuring reliable performance, including a short-circuit withstand time of up to 2.3 µs to provide additional safety margin.
• Lower system cost: Streamlining design processes to reduce system costs and development time.

Gen 4 SiC power modules, discrete components, and bare die are available now through Wolfspeed’s distributor network.

Wolfspeed

Find more datasheets on products like this one at Datasheets.com, searchable by category, part #, description, manufacturer, and more.

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A lightweight runtime security code embedded into a system-on-chip (SoC) for Internet of Things (IoT) applications. That’s the outcome of a collaboration between MediaTek and Italy-based embedded IoT security firm Exein. EE Times’ Editor-in-Chief Nitin Dahad spoke to Gianni Cuozzo, founder and CEO of Exein, to know more about this collaboration that ensures security is an integral part of the development process rather than an afterthought.

Cuozzo, who founded the company in 2018 to address the emerging mandatory cybersecurity regulations, claims it’s the world’s first integration between a chip manufacturer and runtime security software. He also claims it’s the lightest runtime agent available, whether running at the edge or the cloud.

Read the full story on EDN’s sister publication, EE Times.

 

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The post Runtime security code embedded into IoT chip appeared first on EDN.

A friend who was buying an older house was concerned about electrical safety and asked for my opinion as an electrical engineer. All of the AC receptacles (also called outlets) in the house were the two-wire non-grounded type with only a hot (black) and a neutral (white) wire; there were no three-wire receptacles with separate Earth ground (green) as mandated by the National Electrical Code (NEC) in the US since the 1960s, Figure 1. (Other countries have similar requirements, but we’ll stick with the US NEC for this discussion.)

Figure 1 For many decades, home AC-line wiring used a basic two-wire receptacle with hot and neutral wires, but the code was upgraded in the 1960s to mandate a three-wire receptacle with a separate ground wire. Source: NCW Home Inspections

An electrician had told him there were two safety-improvement options: 1) rewire some, or all, of the receptacles to have a true ground, a costly and messy undertaking; or 2) install receptacles with built-in Ground Fault Circuit Interruption (GFCI) functions costing about $20 each at outlets of concern. which is not messy, could be done by anyone with a screwdriver and basic ability, and no electrician needed.

My friend’s questions were these: was using a GFCI on a receptacle without a true ground just a cosmetic, feel-good thing? Did it provide any protection? Full protection? Was it code approved? Most important, would it prevent user shock in case of a fault in the wiring or the load?

My answer was simple: I didn’t know. I assumed you needed a ground for proper GFCI wiring, but the electrical code is complicated with many subtleties.

If you only learned about electricity as part of “Electronics 101” but not from the perspective of the power-electrical code and safety, you’re in for many surprises. There are often requirements that don’t make sense at first, and you are likely to have misconceptions as well. The NEC is very good at what it does and defines, and it characterizes a world which is far different than simply using a qualified AC/DC supply to power your lower-voltage circuits.

I did some research and found that, contrary to my intuition, a GFCI without a formal third-wire ground does provide some protection against some types of faults, but not all. Incidentally, we are talking about a real Earth ground here, not the circuit “common” which is often referred to as “ground” even though it has nothing to do with the Earth ground—a misnomer that is not only widely used but easily leads to sloppy and sometimes dangerous assumptions. Most electronic-circuit “grounds” are not grounds at all, end of story.

A little background: The consumer GFCI was developed in the 1960s; there were earlier designs, but they were subject to false tripping and had higher tripping thresholds. Use of GFCIs was mandated by the NEC since 1968, when it first allowed for GFCIs as a method of protection for underwater swimming pool lights. Throughout the 1970s, GFCI installation requirements were gradually added for 120-volt receptacles in areas prone to possible water contact, including bathrooms, garages, and any receptacles located outdoors. The 1980s saw additional requirements implemented.  During this period, kitchens and basements were added as areas that were required to have GFCIs, as well as boat houses, commercial garages, and indoor pools and spas.  New requirements during the ’90s included crawl spaces, wet bars and rooftops. 

How it works: The operating principle of the GFCI is clear, although implementation has subtleties, of course. The GFCI function is usually built into the AC receptacle and is connected across the three AC-line wires, Figure 2; it is “invisible” to the person doing the installation. Portable and external versions are also available and authorized by the NEC for some situations, but the principle is the same.

Figure 2 Wiring of a GFCI receptacle is the same as for a non-GFCI unit, as the GFCI function is embedded and invisible to the user. Source: PDH Online

In normal operation, current flows between the hot and neutral wires with the load in between the two, and there is no current flow through the ground wire. When there is a fault such as current leaking from one of the active conductor through the load (appliance, tool, hair dryer) and possibly through a user and then to ground—a potential shock situation—the current instead goes directly to ground, as that is the path with far lower impedance than through a person. The safety and shock risk from current flow is reduced to non-dangerous levels.

If there is no ground connection, or the ground wire is defective (thus, a “ground fault”), the user is at risk. The reason is that the fault current no longer has a low-impedance path to ground, and instead goes through the user, Figure 3. At the same time, the current flowing through the hot conductor is not the same as the current returning through the neutral conductor.

Figure 3 If a direct, low-impedance path to ground is absent, fault currents may instead flow through the user to ground, establishing a shock risk. Source: Pressbooks/Douglas College, Canada

This is where the GFCI comes into action: it detects this hot/neutral current imbalance and disconnects the hot and neutral lines from the load. When it senses that imbalance of current, a sensor coil within the GFCI generates a small current that is detected by a sensor circuit. If the sensed current is above a preset threshold, the sensor circuit releases a solenoid, and the current-carrying contacts open (“trip”).

How much imbalance is tolerated? The NEC dictates that residential GFCIs designed to protect people (rather than electrical infrastructure) interrupt the circuit within 25 milliseconds if the leakage current exceeds a range of 4 to 6 milliamps. (The GFCI manufacturer chooses the exact setting.) For equipment-only receptacles, the limit is higher at around 30 milliamps.

Note that GFCIs can’t protect against faults which do not involve an external leakage current, as when current passes directly from one side of the circuit through a victim to the neutral wire. They don’t protect against overloads or short circuits between the hot conductor and neutral.

What about non-grounded GFCIs?: The NEC is an evolving document that is updated every few years to allow new technologies and configurations while disallowing others. GFCI’s provide protection whether or not the house wiring is grounded—that’s why they are called “ground fault” devices and not “shock-protection” ones.

Over the years the NEC has mandated use of grounded GFCIs in new installations, but also formally allowed for retrofit installation without a third-wire ground. In such cases, the three-wire GFCI receptacle or its cover place must be marked “no equipment ground.”

A GFCI will help to protect against electric shock where current flows through a person from a hot or neutral phase to Earth, but it cannot protect against electric shock where current flows through a person from phase to neutral or phase to phase. For example, if someone touches both live and neutral wires the GFCI cannot differentiate between current flows through an intended load versus flows through a person.

When you think about it, not having a third-wire ground at all is the ultimate ground fault. A GFCI does not require an equipment-grounding conductor (green wire) since the GFCI detects an imbalance between the “hot” (black) conductor and the “neutral” (white) conductor.

In short: using a GFCI on a non-grounded receptacle does, indeed, provide some level of protection, even though there is no “ground in which a fault can develop”. The GFCI doesn’t magically produce a ground; it just interrupts power when abnormal current flow is detected. Your electronic devices won’t be protected if there’s a ground fault, for example, and a standard plug-in tester won’t work on the non-grounded GFCI outlet (that can be confusing). Still, an ungrounded GFCI outlet will still shut off in the event of a current-flow fault, so it can help keep users safe.

The answer to the question of using a GFCI in a non-grounded receptacle rather than adding a ground wire is easy: do it. The GFCI provides some protection when the ground wire is faulty, and the absence of a ground wire is certainly a clear fault. It provides some level of protection again user shock under the most common wiring and load failure modes.

Dealing with power-line wiring, faults, regulations, and codes is not trivial, but the rules are based on basic and solid electrical principles. It’s easy to think you understand more about it than you actually do, when you don’t grasp the reasoning behind many of the mandates of the code. While ignorance may be bliss, here it can be dangerous, especially when based on overconfidence or misconceptions.

Bill Schweber is an EE who has written three textbooks, hundreds of technical articles, opinion columns, and product features.

Related Content

• (Under)standing Your Ground
• AC grounding: essential, dangerous, or both?
• AC Grounds: So Essential, Except When They’re Not
• When Poor Grounding Leads to More Grounding
• Mistakes were made, even in a simple 3-wire AC hookup

References

• Angi, “Does a GFCI Outlet Need to be Grounded?”
• com, “406.4(D)(2) Non-Grounding-Type Receptacles”
• NCW Home Inspections, LLC, “3 Prong Grounding Type Receptacles on 2 Wire Ungrounded System: A little history”
• EE World Online, “Basics of ground fault interrupters”
• PDH Online Course E321, “Ground Fault Circuit Interrupters”
• Douglas College/BC/Canada, “7 Electrical Safety: Systems and Devices”
• CPSC Fact Sheet, “What is a GFCI?”
• Harvard University Campus Services, “Ground Fault Circuit Interrupters (GFCI)”
• Inter­national Association of Certified Home Inspectors, “Ground-Fault Circuit Interrupters (GFCIs)”
• The Holmes Group/Make it Right, “Ground Fault Circuit Interrupters (GFCIs)”
• This Old House, “GFCI Receptacles” (video)
• Independent Alliance of the Electrical Industry (IAEI), “GFCI and AFCI Basics”

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After the release of DeepSeek’s R1, a reasoning LLM that matches the performance of OpenAI’s latest o1 model, trade media is abuzz with speculations about the future of artificial intelligence (AI). Has the AI bubble burst? Is it the end of Nvidia’s spectacular AI ride?

EE Time’s Sally Ward-Foxton takes a closer look at the engineering-centric aspects of this talk of the town, explaining how DeepSeek tinkered with AI models as well as interconnect bandwidth and memory footprint. She also provides a detailed account of Nvidia’s chips utilized in this AI head-turner and what it means for Nvidia’s future.

Read the full story at EDN’s sister publication, EE Times.

 

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The post DeepSeek’s AI stunner and the future of Nvidia appeared first on EDN.

This design idea (DI) takes an unusual path to a power-handling DAC by merging an upside-down LM337 regulator with a simple (just one generic chip) PWM circuit to make a 20-V, 1-A current source. It’s suitable for magnet driving, battery charging, and other applications that might benefit from an agile and inexpensive computer-controlled current source. It profits from the accurate internal voltage reference, overload, and thermal protection features of that time proven and famous Bob Pease masterpiece! 

Wow the engineering world with your unique design: Design Ideas Submission Guide

Full throttle (PWM duty factor = 1) current output accuracy is entirely determined by R4’s precision and the ±2% (guaranteed, typically lots better) accuracy of the LM337 internal reference. It’s thus independent of the (sometimes dodgy) precision of logic supplies as basic PWM DACs often are not.

Figure 1 shows the circuit.

Figure 1 LM337 mates with a generic hex inverter to make an inexpensive 1-A PWM current source. (* = 1% metal film)
Iout = 1.07(DF – 0.07), Iout > 0

ACMOS inverters U1b through U1e accept a 10 kHz PWM signal to generate a -50 mV to +1.32 V “ADJ” control signal for the U2 current regulator proportional to the PWM duty factor (DF). Of course, other PWM frequencies and resolutions can be accommodated with the suitable scaling of C1 and C2. See the “K” factor arithmetic below.

DF = 0 drives ADJ > 1.25 V and causes U2 to output the 337’s minimum current (about 5 mA) as shown in Figure 1’s caption.

Iout = 1.07(DF – 0.07)

The 7% zero offset was put in to insure that DF = 0 will solidly shut off U2 despite any possible mismatch between its internal reference and the +5 V rail. It’s always struck me as strange that a negative regulator like the 337 sometimes needs a positive control signal, but in this case it does.

U1a generates an inverse of the PWM signal, providing active ripple cancellation as described in “Cancel PWM DAC ripple with analog subtraction.” Since ripple filter C1 and C2 capacitors are shown sized for 8 bits and a 10-kHz PWM frequency, for this scheme to work properly with different frequency and resolution, the capacitances will need to be multiplied by a factor K:

K = 2(N – 8) (10kHz/Fpwm)
N = bits of PWM resolution
Fpwm = PWM frequency

If more current capability is wanted, the LM337 is rated at 1.5 A. That can be had by simply substituting a heavier-duty power adapter and making R4 = 0.87 ohms. Getting even higher than that limit, however, would require paralleling multiple 337s, each with its own R4 to ensure equal load sharing.

Finally, a word about heat. U2 should be adequately heatsunk as dictated by heat dissipation equal to output current multiplied by the (24 V – Vout) differential.  Up to double-digit wattage is possible, so don’t skimp in the heatsink area. The 337s go into automatic thermal shutdown at junction temperatures above 150oC so U2 will never cook itself. But make sure it will pass the wet-forefinger-sizzle “spit test” anyway so it won’t shut off sometime when you least expect (or want) it to!

Stephen Woodward’s relationship with EDN’s DI column goes back quite a long way. Over 100 submissions have been accepted since his first contribution back in 1974.

 Related Content

• 0 V to -10 V, 1.5 A LM337 PWM power DAC
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• Cancel PWM DAC ripple with analog subtraction but no inverter
• Fast-settling synchronous-PWM-DAC filter has almost no ripple

The post 1-A, 20-V, PWM-controlled current source appeared first on EDN.

A new host bus adapter (HBA) secures all data moving between servers and storage by facilitating quantum-resistant network encryption and real-time ransomware detection in data centers. Broadcom’s Emulex Secure Fibre Channel HBA encrypts all data across all applications while complying with the NIST 800-193 framework, which encompasses secure boot, digitally signed drivers, T10-DIF, and more.

Figure 1 Emulex Secure Fibre Channel HBA provides in-flight encryption with quantum-resistant algorithms. Source: Broadcom

Encryption of mission-critical data is no longer a nice-to-have feature; it’s now a must-have amid the continued rise of ransomware attacks in 2024, costing $5.37 million on average per attack, according to Ponemon Institute’s “Cost of a Data Breach” report. The advent of generative AI and quantum computers further magnifies this risk if data is not encrypted at all points in the data center, including the network.

It’s important to note that data centers have the option of deploying application encryption or network encryption to protect their data. However, network encryption enables real-time ransomware detection while application-based encryption hides ransomware attacks.

Network encryption also offers several important advantages compared to application-based encryption. One is that it preserves storage array services such as dedupe and compression, which are destroyed when using application-based encryption.

Not surprisingly, therefore, IT users are seeking ways to protect themselves against crippling and expensive ransomware attacks; they also want to comply with new government regulations mandating all data be encrypted. That includes the United States’ Commercial National Security Algorithm (CNSA) 2.0, the European Union’s Network and Information Security (NIS) 2, and the Digital Operational Resilience Act (DORA).

These mandates call for enterprises to modernize their IT infrastructures with post-quantum cryptographic algorithms and zero-trust architecture. Broadcom’s Emulex Secure HBA, which secures data between host servers and storage arrays, provides a solution that, once installed, encrypts all data across all applications.

Figure 2 HBA’s session-based encryption is explained with three fundamental tasks. Source: Broadcom

Emulex Secure HBA facilitates in-flight storage area network (SAN) data encryption while complementing existing security technologies. Next, it supports zero-trust platform with Security Protocol and Data Model (SPDM) cryptographic authentication of endpoints as well as silicon root-of-trust authentication.

It runs on existing Fibre Channel infrastructure, and Emulex 32G and 64G Secure HBAs are available in 1, 2, and 4 port configurations. These network encryption solutions offloaded to data center hardware are shipping now.

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The post Host bus adapter boasts quantum-resistant network encryption appeared first on EDN.

Current-mode control LLC considerations

Inductor-inductor-capacitor (LLC) serial resonant circuits, as shown in Figure 1, can achieve both zero voltage switching on the primary side and zero current switching on the secondary side in order to improve efficiency and enable a higher switching frequency. In general, an LLC converter uses direct frequency control, which has only one voltage loop and stabilizes its output voltage by adjusting the switching frequency. An LLC with direct frequency control cannot achieve high bandwidth because there is a double pole in the LLC small-signal transfer function that can vary under different load conditions [1] [2]. When including all of the corner conditions, the compensator design for a direct frequency control LLC becomes tricky and complicated.

Current-mode control can eliminate the double pole with an inner control loop, achieving high bandwidth under all operating conditions with a simple compensator. Hybrid hysteretic control is a method of LLC current-mode control that combines charge control and ramp compensation [3]. This method maintains the good transient performance of charge control, but avoids the related stability issues under no- or light-load conditions by adding slope compensation. The UCC256404 LLC resonant controller from Texas Instruments proves this method’s success.

Figure 1 LLC serial resonant circuits that achieve both zero voltage switching on the primary side and zero current switching on the secondary side. Source: Texas Instruments

Principles of LLC current-mode control

Similar to pulse-width modulation (PWM) converters such as buck and boost, peak current-mode control controls the inductor current in each switching cycle and simplifies the inner control loop into a first-order system. Reference [2] proposes LLC charge control with the resonant capacitor voltage.

In an LLC converter, the resonant tank operates like a swing. The high- and low-side switches are pushing and pulling the voltage on the resonant capacitor: when the high-side switch turns on, the voltage on the resonant capacitor will swing up after the resonant current turns positive; conversely, when the low-side switch turns on, the voltage on the resonant capacitor will swing down after the resonant current turns negative.

Energy flows into the resonant converter when the high-side switch turns on. If you remove the input decoupling capacitor, the power delivered into the resonant tank equals the integration of the product of the input voltage and the input current. If you neglect the dead time, Equation 1 expresses the energy in each switching cycle.

In Equation 1, the input voltage is constant, and the input current equals the absolute of the resonant current. So, you can modify Equation 1 into Equation 2.

Looking at the resonant capacitor, the integration of the resonant current is proportional to the voltage variation on the resonant capacitor (Equation 3).

Equation 4 deduces the energy delivered into the resonant tank.

From Equation 4, it is obvious that the energy delivered in one switching cycle is proportional to the voltage variation on the resonant capacitor when the high-side switch turns on. This is very similar to peak current control in a buck or boost converter, in which the energy is proportional to the peak current of the inductor.

LLC current-mode control controls the energy delivered in each switching cycle by controlling the voltage variation on the resonant capacitor, as shown in Figure 2.

Figure 2 The LLC current-mode control principle that manages the energy delivered in each switching cycle by controlling the voltage variation on the resonant capacitor. Source: Texas Instruments

LLC current-mode control with MCUs

Figure 3 shows the logic of a current-mode LLC implemented with the TMS320F280039C C2000 32-bit microcontroller (MCU) from Texas Instruments, which includes a hardware-based delta voltage of resonant capacitor (ΔVCR) comparison, pulse generation and maximum period limitation [4].

In LLC current-mode control, signal Vc comes from the voltage loop compensator, and signal VCR is the voltage sense of the resonant capacitor. A C2000 comparator subsystem module has an internal ramp generator that can automatically provide downsloped compensation to Vc. You just need to set the initial value of the ramp generator; the digital-to-analog converter (DAC) will provide the downsloped VCR limitation (Vc_ramp) based on the slope setting. The comparator subsystem module compares the analog signal of VCR with the sloped limitation, and generates a trigger event (COMPARE_EVT) to trigger enhanced PWM (ePWM) through the ePWM X-bar.

The action qualifier submodule in ePWM receives the compare event from the comparator subsystem and pulls low the high side of PWM (PWMH) in each switching cycle. The configurable logic block then duplicates the same pulse width to the low side of PWM (PWML) after PWMH turns low. After PWML turns low, the configurable logic block generates a synchronous pulse to reset all of the related modules and resets PWMH to high. The process repeats with a new switching cycle.

Besides the compare actions, the time base submodule limits the maximum pulse width of PWMH and PWML, which determines the minimum switching frequency of the LLC converter. If the compare event hasn’t appeared until the timer counts to the maximum setting, the time base submodule will reset the AQ submodule and pull down PWMH, replacing the compare event action from the comparator subsystem module.

This hardware logic forms the inner VCR variation control, which controls the energy delivered to the resonant tank in each switching cycle. You can then design the outer voltage loop compensator, using the traditional interrupt service routine to calculate and refresh the setting of the VCR variation amplitude to Vc.

For a more detailed description of the hybrid hysteretic control logic, see Reference [1].

Figure 3 LLC current-mode control logic with a C2000 MCU where the signal Vc comes from the voltage loop compensator, and the signal VCR is the voltage sense of the resonant capacitor. Source: Texas Instruments

Experimental results

I tested the current-mode control method described here on a 1-kW half-bridge LLC platform with the TMS320F280039C MCU. Figure 4 shows the Bode plot of the voltage loop under a 400 V input and 42 A load, proving that the LLC can achieve 6 kHz of bandwidth with a 50-degree phase margin.

Figure 4 The Bode plot of a current-mode control LLC with a 400 V input and 42 A load. Source: Texas Instruments

Figure 5 compares the load transient between direct frequency control and hybrid hysteretic control with a 400-V input and a load transient from 10 A to 80 A with a 2.5 A/µs slew rate. As you can see, the hybrid hysteretic control current-mode control method can achieve better a load transient response than a traditional direct frequency control LLC.

For more experimental test data and waveforms, see Reference [5].

Figure 5 Load transient with direct frequency control (a) and hybrid hysteretic control (b), from 10 A to 80 A with a 2.5 A/µs slew rate under a 400 VDC input. Green is the primary current; light blue is the output voltage, with DC coupled; purple is the output voltage, with AC coupled; and dark blue is the output current. Source: Texas Instruments

Digital current-mode controlled LLC

The digital current-mode controlled LLC can achieve higher control bandwidth than direct frequency control and hold very low voltage variation during load transition. In N+1 redundancy and parallel applications, this control method can keep the bus voltage within the regulation range during hot swapping or protecting. So, this control method has been widely adopted in data center power and AI server power with this fast response feature and digital programable ability.

Desheng Guo is a system engineer at Texas Instruments, where he is responsible for developing power solutions as part of the power delivery industrial segment. He has created multiple reference designs and is familiar with AC-DC power supply, digital control, and GaN products. He received a master’s degree from the Harbin Institute of Technology in power electronics in 2007, and previously worked for Huawei Technology and Delta Electronics before joining TI.

Related Content

• Power Tips #84: Think outside the LLC series resonant converter box
• Power Tips #117: Measure your LLC resonant tank before testing at full operating conditions
• Power Tips #122: Overview of a planar transformer used in a 1-kW high-density LLC power module
• Power Tips #97: Shape an LLC-SRC gain curve to meet battery charger needs
• Power Tips #92: High-frequency resonant converter design considerations, Part 2

References

1. Hu, Zhiyuan, Yan-Fei Liu, and Paresh C. Sen. “Bang-Bang Charge Control for LLC Resonant Converters.” Published in IEEE Transactions on Power Electronics 30, no. 2, (February 2015): pp. 1093-1108. doi: 10.1109/TPEL.2014.2313130.
2. McDonald, Brent, and Yalong Li. “A novel LLC resonant controller with best-in-class transient performance and low standby power consumption.” Published in 2018 IEEE Applied Power Electronics Conference and Exposition (APEC), San Antonio, Texas, March 4-8, 2018, pp. 489-493. doi: 10.1109/APEC.2018.8341056.
3. “UCC25640x LLC Resonant Controller with Ultra-Low Audible Noise and Standby Power.” Texas Instruments data sheet, literature No. SLUSD90E, February 2021.
4. Li, Aki, Desheng Guo, Peter Luong, and Chen Jiang. “Digital Control Implementation for Hybrid Hysteretic Control LLC Converter.” Texas Instruments application note, literature No. SPRADJ1A, August 2024.
5. Texas Instruments. n.d. “1-kW, 12-V HHC LLC reference design using C2000 real-time microcontroller.” Texas Instruments reference design No. PMP41081. Accessed Jan. 16, 2025.

 

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The recent design idea (DI) “Negative time-constant and PWM program a versatile ADC front end” disclosed an inventive programmable gain amplifier with integral samples-and-holds. The circuit schematic from the DI appears in Figure 1. Briefly, a PWM signal controls the switches shown. In the X0 positions, a differential signal connected to the inputs of op amps U1a and U1b drives a new voltage sample across capacitor C1 through switches U2a and U2b. Because switch U2c’s X-to-X1 connection is open, capacitor C2 is “holding” a version of the previous sample. This “held” version was amplified by the subcircuit consisting of U2a, U2b, U1c, C1, R1, R2, and R3 with switches in the X1 position. The U1c-based gain-of-two amplifier applies positive feedback through R1 and U2a to the load of C1 in series with the resistance of U2b. This causes the voltage across the load to increase exponentially (with a positive time constant), affording a gain which is a function of the time period that the switches are in the X1 position. The advantage of this approach is that programmable, wideband gains of 60 dB or more can be achieved because the op amps’ maximum closed loop gain is only 6 dB; bandwidth is not sacrificed to achieve a high closed-loop gain.

Wow the engineering world with your unique design: Design Ideas Submission Guide

Figure 1 Two generic chips and five passives make a versatile and unconventional ADC front end.

As with any design, this one has errors and characteristics whose nature must be understood before compensation can be considered. These include passive component tolerances; op amp input currents (negligible) and offset voltages; switch turn-on and turn-off times; leakage currents; as well as resistances and switching-induced charge injections. A non-obvious error can also exist which might be termed a “dead zone”. At time t = 0 when the X1 positions are initially active, the sum of a positive low amplitude VIN – (-VIN) input voltage sample and various errors can yield a negative voltage at the non-inverting input of U1c. Consequentially, U1c’s output voltage would not trend positive and, assuming the analog-to-digital converter (ADC) driven by VOUT accepts only positive voltages, the circuit would not work properly. To understand how to bound this undesirable behavior and for other reasons, it’s wise to develop equations to analyze circuit behavior. Some analytic simplicity is possible when the switches are in the X0 position and operation is mostly intuitive. But operation in the X1 position will require a bit of a deeper dive.

Charge injection error

Op amp input offset voltages and switch resistances are commonly well-understood. As for switch leakage current and charge injection, there’s a referencei that provides an excellent discussion of each. Charge injection Q is most pernicious when the switch transitions from “on” to “off” and a switch terminal is connected to a circuit path which includes a capacitor C and is characterized by a high resistance.

This extends the time for the error voltage Q/C impressed upon the capacitor to “bleed off”. This is not a concern for U2b’s X pin at any time, because both X0 and X1 positions provide a low “on” resistance path. But it must be considered for U2a and U2c when respectively, X0 and X1 turn off. For U2a, X1 is in series with relatively large resistance R1. When U2c’s X1 is turned off and C2 is in hold mode (allowing an analog to digital conversion), C2 sees X1’s multi-megaohm “off” resistance. There is no mechanism to bleed off and recover from U2c’s charge injection error; it is inherent in circuit operation until the PWM reactivates X1 for “tracking” mode, during which time conversions are precluded.

Leakage current

This same high resistance might create a real problem due to leakage current. Such currents flow continuously from U2’s power supplies through its internal ESD diodes to the switch terminals. What saves the circuit from these errors is that an analog to digital (A-to-D) conversion of VOUT can be triggered quickly after X1 turns off, before significant leakage current errors can accumulate. Leakage from terminal X of U2b can be ignored because as mentioned before, it’s always connected to a low resistance through X1 to ground or X0 to U1b’s output. Not so the X terminal of U2a with its connection to moderately high resistance R1. Here the leakage current effect must be considered.

LTspice model and equation validation

Taking all this into account, equations can be developed with reference to the circuit seen in Figure 2. Figure 2 is an illustration of the LTspice file developed to model Figure 1 circuit operation and compare it with the equations (which are also evaluated in the file) to ensure their accuracies.

U2a charge injection is not explicitly shown in the circuit, but is incorporated by summing it with the intended input sample voltage Vin in the .param Vc0 statement. The .param statements and the circuit constitute the model. These statements and the algebraic expressions assigned to voltage sources eq_1 and eq_VOUT validate the equations by allowing direct comparisons with the performance of the model. This is accomplished by graphing simulations of the circuit and evaluations of the voltage sources and confirming that eq_1 = e_1 and that eq_Vout = Vout. Not accounted for are switch turn on and turn off times. Their effects will be addressed later.

Figure 2 The LTspice file comparing the performances of the circuit model and the equations developed of it.

Reference Figure 1 and Figure 2, particularly Figure 2’s .param statements. At time t = 0 when the X0 switches turn off and the X1’s turn on, the voltage across C1 has been initialized to Vc0 as seen in the C1 initial condition (IC) and the .param Vc0 statements. We can write that the current (w) through C1 can be seen in [1].

Therefore:

Assuming a solution of the form:    

Where t = time, [3] and [4] can be seen:

And:

Therefore, the voltage at terminal e_1 can be seen in [6].

To evaluate the voltage at terminal e_2 in the model, it is necessary to convolve the signal at e_1 with the impulse response h(t) of the rc and C2 network shown in [7].

Where the exponential time constant can be seen in [8].

The convolution is given by [9].

This evaluates to [10].

Where:              

Allowing for the U2c charge injection and U1d input offset, shown in [12].

or equivalently:              

The model and this last equation above predict the circuit output at any time t = t1 immediately after the charge injection of U2c has occurred due to the enabling of the X0 switches.

Assessments

Let’s get some worst-case error parameter values for U1 and U2. The original DI proposed specific op amp TLV9164ii, but not a particular 74HC4053. Surprisingly, there are significant differences between parts from different 74HC4053 manufacturers. The MAX74HC4053Aiii looks like a reasonable choice. Let’s consider operation of both IC’s in the commercial temperature range of 0 to 70°C. Refer to Table 1 and Table 2.

Supply	Temperature range	Input current, typical	Input offset voltage	Input offset voltage drift	Open loop gain minimum
5-16 V	-40 to +125oC	± 10 pA	± 1.3 mV	± 0.25 µV/oC	104 dB

Table 1 The TLV9164 maximum parameter values.

Supply	Temperature range	Switch resistance	Switch resistance differences	Flatness, VCOM= ±3V, 0V	COM current	NO current	Charge injection	Switch t(on)	Switch t(off)
± 5 V	0 to 70 oC	125 Ω	12 Ω	15 Ω	± 2.5 nA	± 5 nA	± 10 pC	250 ns	200 ns
5 V	0 to 70 oC	280 Ω	–	–	± 5 nA	± 10 nA	± 10 pC	275 ns	175 ns
3 V	0 to 70 oC	700 Ω	–	–	± 5 nA	± 10 nA	± 10 pC	700 ns	400 ns

Table 2 MAX74HC4053 parameters, maximum values. Note performance degradations when powered by a single supply.

The output of U1c will not move in a positive direction if the sign of the parameter J in [5] is negative. Assuming ADCs whose most negative input value is ground, the circuit will fail to function properly. It’s unlikely that parameters Q1, Voffab, Voffc, and iLeak all take on their worst-case values in a particular circuit, but if they do, Vin will have to be more positive than 10pC/1nF + 2·1.2mV + 1.2mV + 2.5nA * 14300 = 14mV to avoid this “dead zone”. Of course, you’re free to use criterion other than the sum of the worst possibilities, but Caveat Designer!

Another consideration is the circuit settling time in successive PWM periods for the sampling voltage of C1, particularly in the transit between an ADC full-scale voltage to half of its LSB (this is the most extreme case which might not be a requirement for some applications). For a ±5-V powered MAX74HC4053A, two 125-ohm switches in series drive the 1-nF C1. With a 12-bit ADC, the required time is (2·125)·1e-9·ln(212+1) = 2.3 µs. Add the switch on-time of 250 ns, and the PWM should enable the X0 switches for tmin = three of its 1 µs cycles for accurate voltage sample acquisition. By comparison, 8-bit ADC’s can get by with 2 µs.

Calibration

The iLeak·R1 and the temperature-sensitive portions of U1’s input offset voltage errors are negligible in comparison with the ones caused by charge injection. However, as noted in the referencei and in typical curves provided in the switch datasheetiii, the magnitudes and signs of voltages will significantly affect the sizes of the charge injections Q1 (U2a) and Q2 (U2c) and also somewhat the ra, rb and rc resistances. For Q1, ra, and rb, the U1a and U1b input voltages are not determinable from A-to-D conversions. Increasing the values of capacitors C1 and C2 will reduce the Q/C charge injection-created error magnitudes but will also necessitate increases in PWM times. Avoiding such time increases by reducing the R1 value magnifies the errors due to mismatches of ra, rb and rc.

The resistances ra, rb and rc will vary with both temperature and voltage levels. Applying algebra to [13] shows surprisingly that if ra, rb and rc resistances are identical, there is zero error introduced regardless of their value! (This assumes that enough time has elapsed for the e-t/Tc term to be negligible. ) While not identical, the slightly less than 1% maximum mismatch error can be calibrated out, but only for a given set of switch voltages and temperature. The datasheet does not provide the information required to determine the errors that could occur when the voltages and temperature are other than those present during calibration.

With the circuit as it stands, I know of no way to eliminate temperature- and voltage-sensitive errors. But there are errors insensitive to these conditions that can be calibrated out. The following procedure assumes an ADC of negligible error (its resolution and accuracy require further investigation) and conversion factor CF counts/volt, one perhaps present on the assembly line of a product incorporating this design.

For any instance of this circuit to which specific and accurately known Vin and -Vin (See Figure 1) voltages are applied at a given temperature, [13] can be thought of as a function of time and Vin: VOUT(t, Vin). VOUT(t1, Vin), VOUT(t2, Vin), and VOUT(t3, Vin) can be captured such that t3/3 = t2/2 = t1 > tmin. A t1 value of 15 µs is suitable, and Vin = 20 ms avoids the dead zone. (The reason for such a small input voltage is given later.) A t3 of 45 µs applies a gain of less than 24 and keeps things under a 1.8 V full scale A-to-D level. It will be appreciated that et3/T = (et1/T)3 and et2/T = (et1/T)2 and that:

where each VOUT(t, Vin) is scaled by CF to a measurement made by the product line A-to-D. The difference terms cancel the constant terms in [13], and the ratio cancels the A·N term. Such cancellations are necessary if T is to be determined accurately. [14] is a quadratic equation, the desired of the two solutions of which is given by :

From this, the value of T can be obtained (note that T depends slightly on ra because of Req and so it also depends on voltage and temperature):

Further:

Allowing the solution:

N( ) is a function of Vin (see .param N) and is equal to SVin + U where S and U are unknown and again are slightly dependent on the usual. The process of equations [14] to [18] will need to be repeated with a value Vin2 different from Vin to arrive at an AN2 term. We could use different values of t, but we might as well keep the same ones. We can safely choose Vin2 = 25 mV (incurring a charge injection and switch resistances very close to that with Vin = 20 mV) and calculate:

From [13] it can be seen that:

so that:

Given an A-to-D conversion count and reversing [22], it can be seen that:

never forgetting that even this result is influenced by the usual.

Because of their negligible sensitivities to voltage and temperature, NPO/COG capacitors and 25 ppm or better metal film resistors are recommended. 1% or better parts are available at costs around .01 USD in quantity.

Conclusions

This innovative circuit has several features to recommend it. It offers differential to single ended conversion with very high CMRR and wide common mode operating range. It offers gains starting at 6 dB in increments of .6dB limited only by the combination of the sampled voltage and saturation at the positive supply rail. Op-amp gains are no greater than 6 dB, so there is no loss of bandwidth due to operation at high closed-loop gains. But this design has some disadvantages.

A detailed calibration scheme is needed which requires the availability on the production line of an ADC whose requirements for accuracy and resolution have not yet been determined. Even with calibration, various errors which cannot be rescued by calibration can impose an operational “dead zone” for circuit input voltages less than up to 14 mV. Errors due to switch charge injection and switch resistance vary with temperature and applied voltages and are difficult if not impossible to calibrate out. The MAX74HC4053A discussed here is a better ‘4053 than others, but another part may exist with less variations in resistance and charge injection.

I would suggest disconnecting the U2c X0 pin. This connection is of limited usefulness and does damage—it injects charge into U1c, affecting the signal fed through R1 to C1. (The effect on C2 during the “hold” mode can be neglected, being of very short duration due to the X1 rc “on” resistance in the C2 path.) If it is decided to retain this connection, please note that its effects have not been accounted for in the foregoing analysis.

Finally, I’d like to acknowledge the comments of eldercosta, a review and comments with some unique perspectives by David Lundquist, and especially the comments and contributions of Stephen Woodward, who designed the circuit discussed in this DI.

Christopher Paul has worked in various engineering positions in the communications industry for over 40 years.

References

ihttps://www.analog.com/media/en/training-seminars/tutorials/MT-088.pdf

iihttps://www.ti.com/lit/ds/symlink/tlv9164.pdf?ts=1733035616360&ref_url=https%253A%252F%252Fwww.ti.com%252Fproduct%252FTLV9164

iiihttps://www.analog.com/media/en/technical-documentation/data-sheets/MAX4051-MAX4053A.pdf

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The post Error assessment and mitigation of an innovative data acquisition front end appeared first on EDN.

Arm has announced the availability of the first public specification drafted around its Chiplet System Architecture (CSA), a set of system partitioning and chiplet connectivity standards harnessed in a design ecosystem supported by over 60 companies. The companies that are part of the CSA initiative include ADTechnology, Alphawave Semi, AMI, Cadence, Jaguar Micro, Kalray, Rebellions, Siemens, and Synopsys.

CSA aims to offer industry-wide standards and frameworks by facilitating the reuse of specialized chiplets, and thus address the fragmentation caused by compatibility issues in chiplet design. It provides a shared understanding of defining and connecting chiplets while developing composable system-on-chips (SoCs).

A couple of recent announcements show how semiconductor firms are employing CSA to build chiplets as part of Arm Total Design, a platform for deploying chiplet-based compute subsystems. Arm Total Design facilitates custom silicon solutions powered by Arm Neoverse Compute Subsystems (CSS), which includes processor cores, IPs, software, and design tools.

Take the case of Alphawave Semi, which has combined its proprietary I/O dies with an Arm Neoverse CSS-powered chiplet. In this chiplet design, Alphawave Semi has employed AMBA CHI chip-to-chip (C2C) architecture specification to connect artificial intelligence (AI) accelerators.

Figure 1 The block diagram shows a chiplet’s major design building blocks. Source: Alphawave Semi

ADTechnology, another member of Arm’s CSA initiative, has utilized Neoverse CSS V3 technology to create a CPU chiplet for high-performance computing (HPC) and AI/ML training and inference applications. In this chiplet, ADTechnology has incorporated Rebellions’ REBEL AI accelerator; the chiplet design is implemented on Samsung Foundry’s 2-nm gate-all-around (GAA) process node, which has been standardized in the CSA ecosystem.

Figure 2 ADTechnology’s CPU chiplet is powered by Neoverse CSS V3 technology. Source: Arm

AMI, also a member of CSA initiative, is contributing its firmware expertise to accelerate the development of custom chiplet solutions. AMI, the first independent firmware vendor in the Arm Total Design ecosystem, offers modular framework for separating the compute and I/O subsystems. Its pre-configured and pre-tested production quality modules for the reusable chiplets streamline the time and resources required for this stage of development.

A multitude of industry-wide chiplet initiatives—spanning from Arm Total Design to Neoverse CSS to CSA—is a testament to how Arm sees the emergence of chiplets as a major opportunity. It also highlights how Arm sees its place as both chiplet solution provider and ecosystem builder.

The ecosystem-building part is especially crucial because RISC-V companies are also making inroads in the chiplets realm.

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• Imec’s Van den hove: Moving to Chiplets to Extend Moore’s Law

The post Arm’s Chiplet System Architecture eyes ecosystem sweet spot appeared first on EDN.

Today’s high-end system-on-chips (SoCs) rely heavily on sophisticated network-on-chip (NoC) technology to achieve performance and scalability. As the demands of artificial intelligence (AI), high-performance computing (HPC), and other compute-intensive applications continue to evolve, designing the next generation of SoCs will require even smarter and more efficient NoC solutions to meet these challenges.

Although these advancements present exciting opportunities, they also bring significant hurdles. SoC designers face rapid expansion in architecture, time-to-market pressures, scarcity of expertise, suboptimal utilization of resources, and disparate toolchains.

Exponential growth in SoC complexity

SoC designs have reached unprecedented levels of complexity, driven by advancements in process technologies and design tools. Now, SoCs typically include between 50 and 500+ IP blocks, ranging from processor cores and memory controllers to specialized accelerators for AI and graphics.

These blocks, which once contained just tens of thousands of transistors, now house anywhere from 1 million to over 1 billion transistors each. As a result, these SoCs incorporate a staggering total of 1 billion to over 100 billion transistors, reflecting the exponential growth in both scale and sophistication, as shown in the figure below.

The above chart highlights relationship between increasing transistor budgets and use of SIP blocks. Source: Arteris, based on https://rb.gy/qmfcn and https://rb.gy/pgdop

This growth in IP blocks and transistor density has enabled the development of advanced architectures featuring multiple processor clusters. Each cluster typically contains up to 8 or more cores in mainstream designs, with high-performance configurations reaching 32 or more cores.

Today, these clusters are organized into arrays to provide massive parallelism. These cutting-edge designs integrate high-bandwidth memory controllers, dedicated AI accelerators, and sophisticated NoC interconnect systems to ensure seamless communication and scalability.

This unprecedented challenge is manageable by using advanced NoC interconnects, which serve as the backbone for efficient data transfer and communication within the chip. These on-chip networks enable seamless integration of numerous IP blocks. Moreover, high-end SoCs often rely on multiple NoCs, each tailored to specific tasks or subsystems to handle the diverse communication needs across different chip areas.

These NoCs may employ a variety of topologies, depending on the application requirements, such as rings for low-latency communication, trees for hierarchical organization, and meshes for scalability and flexibility.

To address these density and performance challenges, 3D stacking technologies are increasingly being adopted. These approaches integrate multiple layers of logic and memory vertically, enabling higher bandwidth and reduced latency compared to traditional 2D designs.

However, 3D stacking introduces additional complexity in NoC design, such as managing inter-layer communication and thermal constraints, which also require innovative interconnect solutions.

Additional challenges

The increasing sophistication of SoC designs has brought additional challenges driven by the rapid pace of growth in the market. As architectures become more elaborate, designers face mounting pressures to overcome these obstacles and adopt innovative solutions to try to keep pace with industry demands.

These challenges can be summarized as follows:

• Time-to-market pressures: Modern SoC design faces immense competition, where delays can result in significant revenue loss and diminished market share. Traditional methods like manual NoC configuration are time-intensive, often consuming weeks or months, which is unsustainable in fast-paced markets.
• Scarcity of expertise: The growing demand for specialized skills in SoC design outpaces the availability of experienced professionals. Engineering teams are often overburdened, with senior experts spending excessive time on repetitive, manual tasks rather than strategic and high-value design decisions.
• Suboptimal utilization of resources: Manual design methods often result in inefficiencies such as excessive wire lengths, increased power consumption, and physical congestion. These inefficiencies impact the overall performance and escalate both the design complexity and production costs.
• Disparate toolchains: Fragmented workflows in SoC development are a significant bottleneck, with disconnected tools used for floorplanning, connectivity and physical design. The lack of integration across these stages leads to inefficiencies, delays in achieving design closure, and difficulties in maintaining consistency throughout the design process.

Addressing these challenges requires adopting automated design methodologies, enhancing workforce expertise, and integrating toolchains to streamline workflows and reduce inefficiencies.

Designers require smarter NoC solutions

The pressure of this new wave of SoC design complexity is pushing design teams to their limits. An effective approach to managing these challenges is to divide the design into smaller, more manageable pieces by partitioning it into IP blocks.

While this method simplifies individual design tasks, it introduces a new challenge in ensuring seamless integration of these blocks to form a fully functional and optimized SoC. The integration process often reveals unexpected issues, such as mismatched interfaces, timing conflicts and resource contention, which can significantly impact performance and delay time-to-market.

The integration challenges become even more pronounced as SoC designs incorporate increasingly sophisticated components such as AI accelerators and advanced interconnect systems. For instance, the evolution of neural processor units (NPUs) and NoC technologies highlights how rapidly the complexity of SoC architectures has grown.

The first NPUs were typically implemented as arrays of multiply-accumulate (MAC) functions. By comparison, today’s NPUs are far more advanced and may be implemented as arrays of processing elements (PEs), all linked by their own mesh topology NoCs.

Similarly, NoC technology has significantly advanced. First-generation NoCs required manual layout and implementation, including the insertion of pipeline stages. Later generations of NoC technology introduced physical awareness, enabling automatic NoC generation and pipeline stage insertion.

The current generation of NoCs supports higher-end features such as soft tiling. This technology encompasses the automatic replication of processing units (PUs) such as processor clusters in high-level SoCs or PEs in NPUs. It also automatically generates the NoC and configures the network interface unit (NIU) associated with each PU with a unique address.

Features like physical awareness and NoC soft tiling dramatically increase productivity, reduce time to market, and mitigate risk. However, as design complexity continues to grow, additional advancements will be needed to address emerging challenges.

Preparing for the future of SoC design

Successfully realizing next-generation devices is getting harder, especially when it comes to integrating all the IPs into the full SoC. There is a clear and present need for the evolution of tools, including NoC technologies, to address the expanding requirements driven by market shifts such as:

• Automate repetitive and time-consuming tasks, freeing up engineering resources for innovation.
• Accelerate NoC generation without sacrificing performance, power, or quality.
• Adapt to diverse design topologies, seamlessly accommodating both hierarchical and flat NoC structures.
• Optimize across multiple metrics, including wire length, latency and congestion, to deliver high-performing designs that meet tight market windows.
• Empower engineers with user-friendly interfaces and flexible workflows, enabling incremental updates and integration into existing toolchains.

When NoC tools and technologies with these capabilities become available, SoC designers will be able to address these escalating design requirements with greater efficiency and innovation.

In short, next-generation NoC solutions must be engineered to meet today’s challenges while anticipating the accelerating demands of future SoC design.

Andy Nightingale, VP of product management and marketing at Arteris, has over 37 years of experience in the high-tech industry, including 23 years in various engineering and product management positions at Arm.

 

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The post The SoC design: What’s next for NoCs? appeared first on EDN.

Infineon’s first PSOC Control MCUs, based on Arm Cortex-M33 processor, enable secured motor control and power conversion. Supported by Modus Toolbox design tools and software, the entry and mainline devices offer varied performance, features, and memory options.

PSOC Control MCUs—C3M for motor control and C3P for power conversion—can be used in appliances, industrial drives, robots, light EVs, solar systems, and HVAC equipment. Their Cortex-M33 processor runs at up to 180 MHz with a DSP and FPU, while a CORDIC math coprocessor accelerates control loop calculations.

Entry-line MCUs (C3M2, C3P2) feature high-resolution, high-precision ADCs and timers, while mainline MCUs (C3M5, C3P5) add high-resolution PWMs for faster real-time response. The devices are PSA Certified Level 2/EPC2 and include Class B and SIL 2 safety libraries. A crypto accelerator, Arm TrustZone, and secure key storage enable IP protection and firmware updates.

The PSOC Control C3 entry and main lineup comprises 34 devices, all available now.

PSOC Control C3 product page

Infineon Technologies 

Find more datasheets on products like this one at Datasheets.com, searchable by category, part #, description, manufacturer, and more.

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