Збирач потоків

4 months ago I shared the progress of our Open Access Health Tracker from V1 to V2. Today I wanted to share the progress from V2 to V3 on a schematic level. And it's massive. 3x3cm PCB, 3 meters of traces and maxing the capabilities of JLCPCB. Plus an additional 2-3 PCBs for sensors not shown. Next up SLP instead of PCB. submitted by /u/Way_5741 [link] [comments]

submitted by /u/AutoModerator
[link] [comments]

I'm designing a fully-automatic turntable from scratch called the Statimatic STM-01, using: - A Teensy 4.1 - Stepper for tonearm elevation - Stepper for tonearm azimuth movement - Stepper for azimuth clutch - Demultiplexer, to split elevation/azimuth stepper signal - Multiplexer, to handle input buttons (like "play" or "pause") - Shift register, to handle output LED statuses The "turntable" part isn't finished yet (nor is the automatic movement), so I'm just using an AR-XA as the turntable for now. I like records, and I like making stuff, so I decided I wanted to make a turntable. I know it isn't practical, but hey, I'm having fun with it! Please excuse the absolute mess that is the wiring. It is open source, though I'm not sure if GitHub links count as self promotion, so I'll play it safe and leave that out. submitted by /u/BetaMaster64 [link] [comments]

[group 4b][variant 4b] 0x0_ = MISC Opcode Name Description 0x00 NOP No operation 0x01 HLT Halt CPU execution 0x02 INT Trigger software interrupt 0x03 CLC Clear carry flag 0x04 SEC Set carry flag 0x05 CLI Clear interrupt enable flag (disable interrupts) 0x06 STI Set interrupt enable flag (enable interrupts) 0x1_ = MOV Opcode Operands Description 0x10 reg, reg Copy value from register to register 0x11 reg, [mem] Load value from memory address into register 0x12 [mem], reg Store register value to memory address 0x13 reg, #imm Load immediate value into register 0x14 [mem], [mem] Copy value from memory address to memory address 0x15 [mem], #imm Store immediate value to memory address 0x16 reg, [reg] Load value from address held in register (pointer read) 0x17 [reg], reg Store register value to address held in register (pointer write) 0x2_ = ALU Opcode Name Description 0x20 ADD Add register to accumulator 0x21 ADC Add register to accumulator with carry 0x22 SUB Subtract register from accumulator 0x23 SBB Subtract register from accumulator with borrow 0x24 AND Bitwise AND with accumulator 0x25 OR Bitwise OR with accumulator 0x26 XOR Bitwise XOR with accumulator 0x27 NOT Bitwise NOT of accumulator 0x28 CMP Compare (subtract without storing result, sets flags only) 0x29 INC Increment register by 1 0x2A DEC Decrement register by 1 0x2B SHL Shift left, MSB goes to carry, LSB set to 0 0x2C SHR Shift right, LSB goes to carry, MSB set to 0 0x2D ROL Rotate left through carry, MSB goes to carry, carry goes to LSB 0x2E ROR Rotate right through carry, LSB goes to carry, carry goes to MSB 0x3_ = JUMP Opcode Name Description 0x30 JMP Unconditional jump to address 0x31 JZ Jump if zero flag set 0x32 JNZ Jump if zero flag clear 0x33 JC Jump if carry flag set 0x34 JNC Jump if carry flag clear 0x35 JS Jump if sign flag set (result negative) 0x36 JO Jump if overflow flag set 0x4_ = STACK/CALL Opcode Name Description 0x40 PUSH Push register onto stack, decrement SP 0x41 POP Pop value from stack into register, increment SP 0x42 CALL Push PC onto stack, jump to address 0x43 RET Pop PC from stack, return to caller 0x44 IRET Pop PC and flags from stack, return from interrupt handler Github repo with all docs and files: https://github.com/mrFavoslav/8bit-cpu-MESAx8 The 8-bit ALU might not work perfectly at the moment. I recently moved some parts of the design around, so there's a good chance a few wires got messed up or misaligned in the process. Any feedback or bug catching is highly appreciated! I'll be posting my progress here and on https://www.favoslav.cz/blog/ submitted by /u/_Favo_ [link] [comments]

submitted by /u/1Davide [link] [comments]

📰 Газета "Київський політехнік" № 21-22 за 2026 (.pdf) Інформація КП пт, 06/05/2026 - 17:59

A new technology addresses a key performance barrier limiting the use of GaN-on-silicon semiconductors in mainstream RF applications. According to Scott Bibaud, president and CEO of Atomera, this will change the economics of GaN in RF by unlocking breakthrough RF performance on low-cost silicon substrates.

Gallium nitride (GaN) devices for high-performance RF applications are typically built on silicon carbide (SiC) substrates; while they offer robust performance, they are also costly and difficult to scale. On the other hand, silicon substrates offer a lower-cost, more scalable foundation with the potential to support larger wafer sizes and greater compatibility with standard silicon manufacturing.

However, GaN-on-silicon underperforms in RF applications due to parasitic channel losses that reduce efficiency, especially at high frequencies. Enter Atomera’s Mears Silicon Technology (MST), which claims to reduce these losses while offering robust linearity and lower-cost GaN solutions for 5G and other high-frequency RF devices.

MST—a quantum-engineered thin-film technology—introduces a thin, oxygen-modified layer near the surface of the silicon wafer to create a more favorable platform for GaN growth, making silicon a more viable foundation for high-performance RF devices. This controlled layer modifies the silicon lattice structure and helps block the diffusion of electrical dopants. That, in turn, improves crystal quality at the GaN-silicon interface.

MST can improve various wafer-level reliability measures in nitrided oxide planar devices. Source: Atomera

Incize, which provides characterization and modeling services for RF semiconductors, has performed RF characterization of the first MST-enabled samples. The Belgian company reports a substantial reduction in parasitic interface charge and a significant reduction in RF losses.

“Beyond the small-signal improvements, the large-signal results are particularly compelling,” said Mostafa Emam, founder and CEO of Incize. “Then there is a linearity benefit that extends into the high-power regime, approaching performance levels typically associated with advanced RF SOI technologies.”

In Atomera’s own testing, MST enabled more than a 10x reduction in parasitic channel charge, reducing a key mechanism of RF power loss and supporting improved high-frequency GaN device performance. The test data also shows that MST enables devices to handle significant power while maintaining signal quality—linearity—under stress.

Robert Mears, founder and CTO of Atomera, is quick to add that linearity is a top concern for RF designers. “The new data shows MST GaN-on-silicon achieving both the ultra-low RF losses and linearity metrics of advanced trap-rich RF SOI,” he said. “At the benchmark input power of 30 mW, the linearity is exceptional, 1000x better than the GaN-on-silicon reference wafer.”

Atomera, a semiconductor materials and technology licensing company, is based in Los Gatos, California.

Related Content

• GaN on silicon or SiC?
• A Guide to GaN-on-Silicon
• A brief history of gallium nitride (GaN) semiconductors
• Why RF Technologies Should Consider GaN Over Silicon
• GaN-on-Si Technology Makes Headway in RF Applications

The post The RF-ready GaN-on-silicon with lower parasitic losses appeared first on EDN.

The South Wales-based compound semiconductor cluster CSconnected has published its Strength in Places Fund (SIPF) impact report 2026, marking the conclusion of a transformative program that has established South Wales as a globally recognized hub for compound semiconductor innovation...

Next-generation AI and data-center power electronics are placing new demands on switching performance, efficiency and reliability. Meeting those demands requires high-fidelity switching data that reflects real device and module behaviour under realistic operating conditions...

Tiny defects inside a quantum material may hold the key to battery-free electronics powered by energy already floating around us. Credit: AI/ScienceDaily.com Scientists have uncovered a new way to control an unusual quantum phenomenon that could one day help power electronic devices without batteries.

An international research team led by Professor Dongchen Qi from the Queensland University of Technology (QUT) School of Chemistry and Physics and Professor Xiao Renshaw Wang from Nanyang Technological University in Singapore investigated the physics behind the nonlinear Hall effect (NLHE), a quantum phenomenon with significant potential for future energy-harvesting technologies.

Unlike the classical Hall effect, the NLHE can convert alternating electrical signals directly into direct current. This means energy from wireless transmissions or other ambient sources could potentially be transformed into usable electricity without relying on conventional diodes or other bulky electronic components. The NLHE is a sophisticated quantum phenomenon in condensed matter physics where a voltage is generated perpendicular to an applied alternating current, even in the absence of a magnetic field, Professor Qi said.

“This effect allows us to convert alternating signals straight into direct current, which is what’s needed to power electronic devices. In principle, it means sensors or chips that could operate without batteries, drawing energy from their environment.”

Quantum Material Shows Stable Performance at Room Temperature

To better understand how the effect works, the researchers examined a high-quality topological material known for its unusual electronic behavior. Their experiments showed that the nonlinear Hall effect remains stable even at room temperature, an important step toward practical applications outside the laboratory. The team also discovered that temperature plays a key role in determining both the strength and direction of the electrical voltage produced by the material.

How Defects and Atomic Vibrations Control the Effect

At lower temperatures, tiny imperfections within the material had the greatest influence on the quantum effect. As temperatures increased, naturally occurring vibrations in the crystal structure became more important. This shift caused the direction of the generated electrical signal to reverse, revealing a previously unseen mechanism for controlling the phenomenon.

“Once you understand what’s happening inside the material, you can design devices to take advantage of it,” Professor Qi said.

That’s when quantum effects stop being abstract and start becoming useful — supporting future applications ranging from self-powered sensors and wearable technology to ultra-fast components for next-generation wireless networks. The findings provide new insight into how quantum materials behave and could help researchers develop smaller, faster, and more energy-efficient technologies that harvest power from their surroundings.

 

The post Scientists discover a Quantum Effect that Eliminate Batteries appeared first on ELE Times.

Editor’s note: This is a multi-part series on how to design a digital-controlled PFC. Previous entries: 

• How to design a digital-controlled PFC, Part 1
• How to design a digital-controlled PFC, Part 2
• How to design a digital-controlled PFC, Part 3

High efficiency is a mandatory requirement in some applications, especially in data centers. The recently announced 80 Plus Ruby certification sets the highest efficiency standard for data center power-supply units (PSUs), as shown in Table 1. The new efficiency requirement is not only higher than 80 Plus Titanium at each load condition, but also requires 90% efficiency at a 5% load, which has never been specified before.

 

 

 

80 Plus test type	230V internal redundant
Percentage of rated load	5%	10%	20%	50%	100%
80 Plus Titanium		90%	94%	96%	91%
80 Plus Ruby	90%	91%	95%	96.5%	92%

Table 1 “Ruby” is the most recent and most stringent of the 80 Plus certification levels

With totem-pole bridgeless power factor correction (PFC) offering the best efficiency among all PFC topologies, digital control can further push the efficiency capabilities of this topology to new levels. In the fourth and final installment of this series, I will first introduce several digital methods to improve efficiency and then discuss some special PFC requirements including re-rush current control, electrical metering (e-metering) and PFC with a baby boost converter.

Dynamic dead time to achieve ZVS for synchronous switch

Theoretically, the PFC synchronous switch can operate with zero voltage switching (ZVS), but there must be a proper dead time between when the boost switch turns off and the synchronous switch turns on. As illustrated in Figure 1, assuming a positive cycle, when boost switch Q2 turns off, the inductor current (IL) starts to charge the output capacitance (COSS) of Q2 and discharge the output capacitance COSS of Q1, and the switch-node voltage rises.

If Q1 turns on before the switch-node voltage rises to the output voltage (VOUT), this is hard switching, and the switching losses are high. If Q1 turns on too late after the switch-node voltage rises to VOUT, the current will conduct in the third quadrant of Q1 with diode-like behavior. Since the gallium nitride field-effect transistor used for Q1 has a higher VSD drop compared to a silicon metal-oxide semiconductor field-effect transistor body diode, this induces a higher third-quadrant conduction loss.

Figure 1 This equivalent circuit describes a PFC synchronous switch during dead time. (Source: Texas Instruments)

Ideally, Q1 should turn on at the exact moment when the switch-node voltage rises to VOUT. Given the IL, VOUT and COSS of Q1 and Q2, the following equation calculates the time to charge the switch node from 0 to VOUT:

You can use firmware to dynamically adjust the dead time calculated from the equation to maintain ZVS for the synchronous switch.

CCM_TCM multimode control

A totem-pole bridgeless PFC can operate in either continuous conduction mode (CCM) or triangular current mode (TCM); each has its advantages and disadvantages. Table 2 provides a high-level comparison between the two modes.

	CCM operation	TCM operation
Pros	Low peak-to-peak IL ripple. Simple control.	ZVS.
Cons	Hard switching – high switching losses.	High peak-to-peak IL ripple. Requires multiphase interleaved operation to reduce current ripple for high-power applications, resulting in low power density and high costs. Complex control.

Table 2 Continuous conduction mode (CCM) and triangular current mode (TCM) options both have pros and cons for totem-pole power factor correction (PFC) operation purposes.

Ideally, the totem-pole bridgeless PFC could operate with multimode, as shown in Figure 2. At heavy loads or at the peak of an AC half cycle, the desired PFC input current is high and the PFC operates in CCM mode. When the load reduces or around the AC zero-crossing area where the desired PFC input current is low, the PFC switches to TCM mode and operates with ZVS.

Compared to pure CCM mode, this multimode operation has better efficiency at light loads because of ZVS. Compared to pure TCM mode, because the inductor current ripple is much lower, there is no need to use multiphase interleaved operation; therefore, this multimode operation significantly reduces the size and system costs. By combining the advantages of both CCM and TCM, this multimode operation can meet both high-efficiency and high-power-density requirements.

Figure 2 CCM_TCM multimode operation can meet both high-efficiency and high-power-density requirements. (Source: Texas Instruments)

Reference 1 provides details about this control method and its implementation. Figure 3 compares the efficiency (tested on the same board) between this CCM_TCM multimode control method and traditional CCM control, with efficiency improving as much as 2%.

(a)	(b)

Figure 3 CCM_TCM multimode control delivers efficiency improvements versus traditional CCM control in both low line (a) and high line (b) environments. (Source: Texas Instruments)

Special burst mode – AC cycle skipping

Burst mode is widely used to improve efficiency at light loads. Unlike traditional pulse-width modulation (PWM) pulse-skipping burst mode, where you skip PWM pulses randomly, here I would like to introduce a special burst mode: AC cycle skipping, which is you skip one or more AC cycles in light loads.

In other words, you would turn the PFC off for one or more AC cycles and turn the PFC back on for the next AC cycle. The turnon and turnoff instance occurs at the AC zero crossing such that the whole AC cycle is skipped. Since PFC turnon and turnoff at inductor current equal zero, there is less stress and electromagnetic interference.

The number of AC cycles to skip is reverse-proportional to the load; the lighter the load, the more AC cycles skipped. Figure 4 shows the skipping of one and two AC cycles, respectively. Channel 1 is the AC voltage, and channel 4 is the AC current.

(a)	(b)

Figure 4 Shown here is AC cycle skipping at a light loads: one cycle (a) and two cycles (b). (Source: Texas Instruments)

Once the PFC turns off, the switching losses, driving losses and reverse-recovery losses all drop to zero, and the power losses are just the PFC standby power.

When turning off the PFC to skip AC cycles, both the current loop and voltage loop need to be frozen; otherwise, the integrators in those loops will build up to generate a big PWM pulse when the PFC turns back on, causing a large current spike.

Determining whether the PFC enters a light load requires the load information. Normally there is no current sensor at the PFC output; therefore, it’s not possible to directly measure the output load. However, because the PFC voltage-loop output is proportional to the load, you can use the voltage-loop output as a rough indicator to determine whether the PFC is operating with a light load.

If you must precisely skip an appropriate number of AC cycles to maintain VOUT ripple within a specified range, you will need accurate load information, which you can obtain through an integrated e-meter function that I will discuss after the next section.

A big concern with AC cycle skipping is the VOUT drop during a load transient. Assuming that a load step-up occurs when the PFC is off, VOUT may drop too much.

To address this issue, you can compare VOUT to a predefined threshold through a comparator. Once VOUT is below this threshold, the PFC will immediately exit burst mode, disable AC cycle skipping, and return to normal operation. The PFC will handle the transient response as if there is no such special burst mode.

AC cycle skipping can also help reduce total harmonic distortion (THD) at light loads. Reference 2 compares THD with and without this method.

Re-rush current limit

The AC input voltage could suddenly drop out when PFC is operating normally. Since the load is still applied, the PFC VOUT could drop to a lower value. Then, when the AC voltage returns, if the AC input voltage is higher than VOUT, there will be an inrush current. This current is called the re-rush current.

Previously, the re-rush current was unspecified and there was no special control action for this event, it solely relied on the power-stage components’ ability to handle re-rush current. Test results show that re-rush current can jump more than 10 times higher than the PFC-rated maximum input current. Such a high re-rush current can either damage the power supply or reduce its lifetime.

The recently released Modular Hardware System– Common Redundant Power Supply (M-CRPS) specification requires limiting re-rush current when the input voltage resumes after an input brownout or blackout event on the power supply used in a data center. As shown in Figure 5, the root-mean-square (RMS) value of re-rush current should not exceed 5 times the maximum PSU rating over one-half cycle of input frequency, or 3.5 times the maximum PSU rating over one cycle of input frequency. In addition, the input current of the PSU should settle to a value less than or equal to two times the maximum PSU rating of the PSU within two cycles of the input frequency after applying the AC input.

Figure 5 The Modular Hardware System– Common Redundant Power Supply (M-CRPS) specification documents limits on both re-rush current and timing. (Source: Texas Instruments)

Reference 3 provides a firmware-based solution to handle this re-rush current so that when the AC voltage comes back from dropout, both the re-rush current (when VIN > VOUT) and the non-re-rush current (when VIN < VOUT) are well controlled – not exceeding the M-CRPS limit specification, but high enough to rapidly boost VOUT.

E-metering

Power supplies in data centers are required to measure the input power in real time and report the measurement to the host; this is called e-metering. The M-CRPS specification requires an input power measurement error within ±1% when the load is >125W, within ±1.25W when the load is between 50W and 125W, and within ±5W when the load is <50W. To achieve such high measurement accuracy, the e-meter function is traditionally implemented through a dedicated metering device, as shown in Figure 6a.

(a)	(b)

Figure 6 These circuit diagrams show a traditional e-meter and PFC control (a), as well as combining an e-meter with PFC control (b). (Source: Texas Instruments)

A current shunt placed on the PFC input side senses the input current, with a voltage divider (not shown in Figure 6a) across the AC line and AC neutral senses the input voltage. A dedicated metering device receives this current and voltage information and calculates the input power and input RMS current information, sending the results to the host.

With a digital controller, since analog-to-digital converters (ADCs) of the microcontroller (MCU) are measuring both the input voltage and input current, it becomes possible to integrate the e-meter function into PFC control code. Figure 6b shows this e-meter configuration.

A current shunt senses the input current and an isolated delta-sigma modulator (the AMC1306 from Texas Instruments) measures the voltage drop across the current shunt. The delta-sigma modulator output is sent to the PFC controller MCU. The current information will be used for both e-metering and PFC current-loop control. A voltage divider senses the input voltage, which is then measured by the MCU’s ADC directly, just as in traditional PFC control. Reference 4 has more details about e-meter implementation and calculation.

Integrating e-meter functionality into PFC control code eliminates the need for a dedicated metering device, not only reducing system costs, but also simplifying printed circuit board layout and expediting the design process.

PFC with a baby boost converter

In server applications, a bulk capacitor (CBULK in Figure 7) is required to hold PSU output in regulation for more than 10mS after AC dropout. To accomplish this, a 3kW server PSU would need a total capacitance of over 1.3mF, which would consume at least 30% of the overall space. To improve power density, you must reduce the bulk capacitance.

Adding a baby boost converter between PFC and DC/DC, as shown in Figure 7 and described in Reference 5, can achieve high power density. The baby boost converter is a compact boost converter that only operates during AC dropout events.

Figure 7 A PFC with a baby boost converter can achieve high power density. (Source: Texas Instruments)

Figure 8 is a flow chart of baby boost converter operation. During normal operation, the baby boost converter is off and bypassed by a BYPASS FET Q4. When AC line dropout occurs and VBULK drops to a certain level, Q4 turns off, and the baby boost converter turns on to allow VBB to maintain its nominal value. If AC power returns, VBULK will rise; once VBULK rises to a certain level, MCU turns off the baby boost converter, turns on BYPASS FET Q4, and the PFC resumes normal operation.

Figure 8 This flow chart outlines the various stages of baby boost converter operation.

Conclusion

I hope that the information imparted in this series enables you to design your own digital-controlled PFC and meet ever-more-strict specifications. You will find that digital control is so flexible that is possible to implement advanced control algorithms that would be difficult to implement with analog control. A digital-controlled power supply also offers impressive performance.

References

1. Sun, Bosheng. “A novel CCM-TCM multimode control method for totem-pole bridgeless PFC.” Texas Instruments Analog Design Journal article, literature No. SLYT877, 1Q 2026.
2. Sun, Bosheng. “AC cycle skipping improves PFC light-load efficiency.” Texas Instruments Analog Design Journal article, literature No. SLYT585, 3Q 2014.
3. Sun, Bosheng. “How to limit PFC re-rush current.” Texas Instruments Analog Design Journal article, literature No. SLYT865, 1Q 2025.
4. Sun, Bosheng. “A low-cost and high-accuracy e-meter solution.” EDN, Aug. 26, 2024.
5. Yu, Sheng-Yang, Benjamin Genereaux, and LiehChung Yin. “Improve power density with a baby boost converter in a PFC circuit.” Texas Instruments Analog Design Journal article, literature No. SLYT830, 2Q 2022.

Related Content

• How to design a digital-controlled PFC, Part 1
• How to design a digital-controlled PFC, Part 2
• How to design a digital-controlled PFC, Part 3
• A low-cost and high-accuracy e-meter solution

The post How to design a digital-controlled PFC, Part 4 appeared first on EDN.