Збирач потоків

I wanted to start a small side project to 'calm down' from my last big one. And because we need something to replace the old FM radio in our dining room, I started building this, an active smart speaker. Currently, I have what you can see on the pictures. I pulled all the parts out of an old multimedia system my mother gave to me. The donor-board can be seen on one of the pictures. It was to big to fit in the speakers I am planing to use, so I decided to split it up by transferring the components onto a self soldered PCB and throwing out unnecessary parts out in the process. The amplifier board was quite easy and done in a few hours, but the power supply took quite long as I paid careful attention while building it because you know, things plugged into the mains. The PSU originally put out multiple voltages for not only the Amp itself(24V) but also the Vacuum fluorescent display and other shenanigans I don't need. I threw out everything except the 24V for the amplifier and the 5V rail to power the RPI and micro-controller that I will put into the device for the 'smart' part. I still have to isolate the bottom of the Power supply and build a small shielding for it to eliminate noise as it will be sitting directly behind the amplifier part. Like I said, apart from these two PCBs I will also be putting a RPI1 and a STM32 with a LCD screen and rotary encoder into this thing to give it streaming capabilities. I will keep you up to date on the progress! submitted by /u/FloTec09 [link] [comments]

КПІ ім. Ігоря Сікорського на брифінгу «EcoEducation: експертиза для екоосвіти нової генерації» kpi ср, 10/08/2025 - 20:29

❤️ Запрошуємо на благодійний концерт японського піаніста Темпея Накамури kpi ср, 10/08/2025 - 20:15

There’s been interest recently here in the land of Design Ideas (DIs) in a family of simple interface circuits for pulse width modulation (PWM) control of generic voltage regulators (both linear and switching). Members of the family rely on the regulator’s internal voltage reference and a discrete FET connected in series with the regulator’s programming voltage divider. 

PWM uses the FET as a switch to modulate the bottom resistor (R1) of the divider, so that the 0 to 100% PWM duty factor (DF) varies the time-averaged effective conductance of R1 from 0 to 100% of its nominal value.  This variation programs the regulator output from Vo = Vs (its feedback pin reference voltage) at DF = 0 to Vo = Vs(R2/R1 + 1) at DF = 100%.

Wow the engineering world with your unique design: Design Ideas Submission Guide

Some of these circuits establish a linear functionality between DF and Vo. Figure 1 is an example of that genre as described in “PWM buck regulator interface generalized design equations.”

Figure 1 PWM programs Vo linearly where Vo = Vs(R2/(R1/DF) + 1).

For others, like Figure 2’s concept designed by frequent contributor Christopher Paul and explained in “Improve PWM controller-induced ripple in voltage regulators”…it’s nonlinear…

Figure 2 PWM programs Vo nonlinearly where Vo = Vs(R2/(R1a/DF + R1b + R1c) + 1).

Note that for clarity, Figure 2 does not include many exciting details of Paul’s innovative design. See his article at the link for the whole story.

The nonlinearity problem

However, to explore the implications of Figure 2’s nonlinearity a bit further, in the example of the circuit provided in Paul’ DI:  

R1a = 2490 Ω
R1b = 2490 Ω
R1c = 4990 Ω
Vs =  0.800 V
R2 = 53600 Ω

Which, if we assume 8-bit PWM resolution, provides the response curves shown in Figure 3.

Figure 3 The 8-bit PWM setting versus DF = X/255. The left axis (blue curve) is Vo = 0.8(53600/(2490/(X/255) + 7480) + 1). The right axis (red curve) is Vo volts increment per PWM least significant bit (LSBit) increment.

Paul says of this nonlinear response: “Although the output voltage is no longer a linear function of the PWM duty cycle, a simple software-based lookup table renders this a mere inconvenience. (Yup, ‘we can fix it in software!’)”Of course, he’s absolutely right: For any chosen Vo, a corresponding DF can be easily calculated and stored in a small (256-entry) lookup table. 

However, translating from the computed DF to an integer 8-bit PWM code is a different matter. Figure 3’s increment-vs-increment red curve provides an important caveat to Paul’s otherwise accurate statement.

If the conversion from 8-bit 0 to 255 code to the 0.8 V to 5.1 V, or 4.3V Vo span, were linear, then each LSBit increment would bump Vo by a constant 15.8 mV (= 4.3 V/256). But it isn’t. 

And, as Figure 3’s red curve shows, due to the strong nonlinearity of the conversion, the 8-bit resolution criterion is exceeded for all PWM codes < 75 and Vo < 3.77 V = 74% of full scale. 

And it gets worse: For Vo values down near Vs = 0.8 V, the LSBit increment soars to 67 mV (= 4.3 V/64). This, therefore, equates to a resolution of not 8 bits, but barely 6.

The fix

Unfortunately, there’s very little any software fix can do about that. Which might make nonlinearity for some applications perhaps more than just an “inconvenience?” So what could fix it?

The nonlinearity basically arises from the fact that only a fraction (R1a) of the total R1abc resistance is modulated by PWM, as the PWM DF changes, that fraction changes, which in turn changes the rate of change of Vo versus DF. In fact, it changes this by quite a lot.

Getting to specifics, in the example of Paul’s circuit provided in his DI, we see they make the modulated resistance R1a only 25% of the total R1 resistance at DF = 100%, with this proportion increasing to 100% as DF goes to 0%. This is obviously a big change concentrated toward lower DF.

A clue to a possible (at least partial) fix is found back in the observation that the nonlinearity and resolution loss originally arose from the fact that only a small fraction (25% R1a) of the total R1abc resistance is modulated by PWM. So, perhaps a bigger R1a fraction of R1abc could recover some of the lost resolution. 

As an experiment, I changed Paul’s R1 resistor values to the following.

 R1a = 7960 Ω
R1b = 1000 Ω
R1c = 1000 Ω

This makes R1a now 80% of R1abc instead of only 25%. Figure 4 illustrates the effect on the response curves. 

Figure 4 The impact of making R1a 80% of R1abc. The left axis (blue curve) is Vo = 0.8(53600/(7960/(X/255) + 2000) + 1). The right axis (red curve) is Vo volts increment per PWM LSBit increment.

Figure 4’s blue Vo versus PWM curve is obviously still nonlinear, but significantly less so. But perhaps the more important improvement is to the red curve: Unlike the previous erosion of resolution at the left end of the curve to 67 mV per PWM LSBit to just 6 bits, Figure 3 maxes out at 21 mV, or 7.7 bits.

Is this a “fix?” Well, obviously, 7.7 bits is better than 6 bits, but it’s still not 8 bits, so resolution recovery isn’t perfect. Also, my arbitrary shuffling of  R1 ratios is almost certain to adversely impact the spectacular ripple attenuation cited in Christopher Paul’s original article. Mid-frequency loop gain may also suffer from the heavier loading on C2 and R2 imposed by the reduced R1c value. This could lead to a possible deterioration of the transient response and noise rejection. Perhaps C2 could be increased to moderate that effect.

Still, it would be fair to call it a start at a fix for nonlinearity that lay beyond the reach of software.

Stephen Woodward’s relationship with EDN’s DI column goes back quite a long way. Over 100 submissions have been accepted since his first contribution back in 1974.

Related Content

• Improve PWM controller-induced ripple in voltage regulators
• PWM buck regulator interface generalized design equations
• A nice, simple, and reasonably accurate PWM-driven 16-bit DAC
• Brute force mitigation of PWM Vdd and ground “saturation” errors

The post PWM nonlinearity that software can’t fix appeared first on EDN.

How do isolated comparators differ from standard comparators? What are their primary applications in analog and power electronics? Here is a brief review of this critical building block and what design engineers need to understand about its application. The article also presents a few popular isolated comparators and what makes them suitable for specific designs.

Read the full article at EDN’s sister publication, Planet Analog.

Related Content

• Designing with comparators
• Comparators Rival Respect of Sibling Amps
• Understanding the analog voltage comparator
• Understanding precision comparator applications
• Hysteresis–Understanding more about the analog voltage comparator

The post A closer look at isolated comparators appeared first on EDN.

Optoelectronic component maker Optrans Corp of Kawasaki City, Japan, in partnership with Marktech Optoelectronics Inc of Latham, NY, USA (a joint venture between Optrans and Marktech International that designs and manufactures optoelectronics components and assemblies), has released new ultraviolet and near-ultraviolet LEDs emitting at 365nm, 395nm and 405nm...

Chinese automotive semiconductor supplier NOVOSENSE Microelectronics, China’s largest tier-1 automotive supplier United Automotive Electronic Systems Co Ltd (UAES, a joint venture established in 1995 by China’s Zhonglian Automotive Electronics Co Ltd and Germany’s Robert Bosch GmbH) and InnoScience (Suzhou) Technology Holding Co Ltd — which manufactures gallium nitride (GaN) power chips on 8” silicon wafers — have signed a strategic cooperation agreement to jointly advance power electronics for new energy vehicles (NEVs)...

OMNIVISION announced its latest-generation automotive image sensor: the OX08D20, 8-megapixel (MP) CMOS image sensor with TheiaCel technology.

This new device is an upgrade to the popular OX08D10 sensor for exterior cameras used in advanced driver assistance systems (ADAS) and autonomous driving (AD). The device will be introduced at AutoSens Europe.

“OMNIVISION’s OX08D10 image sensor with TheiaCel technology was launched at AutoSens Brussels in 2023 and has been a highly successful product for automotive OEMs seeking a single device that integrates all of the most important image sensor features, including best-in-class low-light performance, LED flicker mitigation, compact size, superior image quality at high temperatures, and low power,” said Dr. Paul Wu, director of automotive product marketing, OMNIVISION. “Our mission is to partner with our customers to solve their toughest challenges. That’s why our latest-generation devices are designed with innovative features our customers need.”

The OX08D20 features all of the benefits of the OX08D10, as well as:

• An innovative capture scheme developed in collaboration with Mobileye that significantly reduces the motion blur of nearby objects (while driving) and improves low-light performance
• An upgrade to 60 frames per second (fps) to enable dual-use cameras
• Updated cybersecurity to match the latest industry standard, MIPI CSE 2.0

In addition to industry-leading low-light performance, the device has low power consumption and comes in an a-CSP package that is 50% smaller than other exterior sensors in its class.

The post OMNIVISION Introduces Next-Generation 8-MP Image Sensor For Exterior Automotive Cameras appeared first on ELE Times.

OMNIVISION launched the latest addition to its popular Nyxel near-infrared (NIR) technology family: the new OX05C—the automotive industry’s first and only 5-megapixel (MP) back-side illuminated (BSI) global shutter (GS) high dynamic range (HDR) sensor for in-cabin driver and occupant monitoring systems (DMS and OMS). The OX05C’s exceptional capabilities will be demonstrated at AutoSens Europe.

The OX05C GS HDR sensor delivers extremely clear images of the entire cabin, enabling improved algorithm accuracy even in challenging high-brightness lighting conditions. It features a pixel size of just 2.2µm and OMNIVISION’s revolutionary Nyxel NIR technology, which achieves the world’s class-leading quantum efficiency (QE) at the 940nm NIR wavelength to further improve DMS and OMS capabilities in low-light conditions. The OX05C has on-chip RGB-IR separation, relieving the burden of a dedicated image signal processor (ISP) and backend processing, thus freeing extra bandwidth for other tasks.

At just 6.61mm x 5.34mm, the OX05C1S package is 30% smaller than its predecessor, the OX05B (7.94mm x 6.34mm), providing automotive OEMs with greater design flexibility to place the camera in various locations within the cabin. Moreover, OEMs can use the same camera lens when upgrading from the OX05B to the newer OX05C, providing a design and cost advantage.

“We have applied OMNIVISION’s unparalleled HDR technology and top-notch RGB-IR processing algorithm for security cameras to the automotive market. This is a significant step forward in the advancement of DMS and OMS cameras,” said Dr. Paul Wu, Director of automotive product marketing at OMNIVISION. “Our OX05C sensor has industry-leading perception performance. Even in bright light conditions, the driver’s face is not washed out, and the camera can detect distractions or drowsiness.” Added Wu, “DMS cameras are a critical safety component for ADAS (advanced driver assistance systems), which is why OMNIVISION strives to continually improve them.”

“Global shutter combined with HDR and Nyxel NIR sensitivity makes the OX05C a real step forward for in-cabin imaging,” said Detlef Wilke, Vice President Innovations & Strategic Partnerships at Smart Eye. “It gives our algorithms the consistency they need to track driver attention and occupant status in everything from bright sunlight to near darkness, while the smaller form factor offers OEMs more flexibility in camera placement.”

Compared to rolling-shutter HDR sensors, the GS HDR OX05C avoids interference from other IR light sources in the cabin, greatly improving the RGB image quality and providing flexibility to enable more capture scheme and functions in real applications. The OX05C features integrated cybersecurity and supports simultaneous driver and occupant monitoring with a single camera, reducing complexity, cost and space. In addition, OMNIVISION developed the sensor with strong ecosystem partner support to provide a complete, seamless solution for automotive OEMs.

The post OMNIVISION Announces Automotive Industry’s First Global Shutter HDR Sensor for In-Cabin Driver and Occupant Monitoring Systems appeared first on ELE Times.

Vishay Intertechnology, Inc. announced that it has successfully delivered on the ambitious expansion of its inductor and frequency control device (FCD) product lines announced in September 2024, significantly increasing the breadth and availability of components now in the field. The company has released more than 2000 new SKUs across nearly 100 series across inductors and frequency control devices, with continued rollouts underway in 2025.

The expanded offering simplifies sourcing for Vishay customers and supports more applications through broader inductance and voltage ranges, improved noise suppression, and additional size variations to fit even the smallest PCB footprints. Recent launches include new wireless charging inductors, common-mode chokes, high current ferrite impedance beads, and TLVR inductors, as well as nearly 15 new FCD products.

“This expansion was designed to deliver on our commitment to give customers maximum design flexibility, and the market response has confirmed that we’ve hit the mark,” said Mike Husman, Senior Vice President, Inductor Division, at Vishay. “We’re now seeing the impact of this expansion in the field — thousands of new SKUs, strong uptake through distribution, and a clear signal from our customers that they value the depth and readiness of our offering.”

To support this growth, Vishay continues to invest in global production capacity, including expansions in Asia, Mexico and the Dominican Republic. In response to the industry’s increasing demand for diversified manufacturing locations — and part of Vishay’s strategy of vertically integrated, resilient manufacturing — flagship Vishay-produced IHLP power inductors are now shipping from the company’s La Laguna plant in Gómez Palacio, Durango.

The momentum continues in 2025, with more product series set to launch in the coming months. In total, the company expects to exceed 3000 new SKUs across inductors and frequency control devices from its overall expansion effort, supporting increased design-in activity across industrial, telecom, and consumer applications.

The post Vishay Intertechnology Expands Inductor Portfolio with 2000+ New SKUs and Increased Capacity appeared first on ELE Times.

Keysight Technologies will demonstrate 20 advanced AI-enabled 6G and wireless solutions designed to accelerate 6G and wireless innovation at India Mobile Congress 2025. The solutions on display include network digital twins, AI-driven channel modeling, and high-fidelity emulation, which enable rapid validation of beamforming, interference mitigation, and ultra-low latency, advancing next-generation 6G and wireless communication systems. In addition, Prasad Petkar, Keysight India Wireless SEO Manager, will present a paper entitled, “Engineering the 6G Future: Accelerating Wireless Innovation from Lab to Live Networks.”

Keysight Technologies will showcase its cutting-edge AI-enabled 6G and wireless solutions from October 8–11, 2025, at Booth No. C10 in Yashobhoomi, New Delhi.

Keysight will feature the following demonstrations:

• mMIMO Design from FR1 to FR3 / New 6G Frequency – Sub THz and FR3: Keysight will show how it’s enabling the leap to 6G with advanced mMIMO design across FR1 to FR3, including new sub-THz bands. These next-generation test solutions unlock faster speeds, wider bandwidths, and reliable performance, helping accelerate future wireless technologies from concept to real-world deployment.
• 5G Interference Mitigation: The demo includes advanced spectrum monitoring, signal analysis, and emulation workflows to detect and isolate interfering signals across licensed and unlicensed bands. Leveraging real-time visibility and AI-driven analytics, the solution accelerates root-cause identification, optimizes spectrum usage, and validates network robustness under diverse interference scenarios critical for 5G evolution.
• NTN Design Optimization: This demo showcases advanced modeling and test workflows for satellite-to-terrestrial integration, addressing propagation delays, Doppler effects, and dynamic channel conditions. By combining link-level validation, system simulation, and real-world emulation, the solution enables engineers to optimize performance, spectrum efficiency, and interoperability for emerging 5G-advanced and 6G NTN deployments.
• RF Environment Emulation: The demo recreates real-world wireless conditions in the lab and highlights how innovators can test devices, networks, and use cases under complex RF scenarios including mobility, interference, and multi-path. This ensures reliable performance, faster validation, and reduced field testing costs for 5G and 6G.
• Device Security Defense: The demo integrates protocol fuzzing, penetration testing, and threat emulation to expose vulnerabilities across wireless, application, and cloud layers. Using automation and real-world attack libraries, the solution validates device resilience against cyberattacks, enabling engineers to harden designs, ensure standards compliance, and accelerate secure product deployment.
• Digital Twins Emulation: This demo enables accurate, scalable modeling of devices and networks under real-world conditions. By combining system simulation, AI-driven analytics, and RF environment emulation, the solution provides end-to-end validation of designs and architectures, helping engineers optimize performance and reliability across 5G-advanced and 6G.

The post Keysight to Demonstrate AI-enabled 6G and Wireless Technologies at India Mobile Congress 2025 appeared first on ELE Times.

I wanted to share a common issue with the LM358 that might help others troubleshooting similar problems. The Problem (Left Circuit) Built a simple non-inverting amplifier (gain ≈ 4.9) using an LM358 with ±9V rails. The output showed significant crossover distortion around zero-crossing - you can see the characteristic "flattening" in the waveform. Root Cause The LM358 uses an NPN output stage that pulls high well but relies on an internal current source to pull low. When driving high-impedance loads (like a scope probe directly), there's insufficient sink current to rapidly transition through zero, creating a dead zone. A Solution (Right Circuit) Adding a 1kΩ pull-down resistor (RL) from output to the negative rail (-9V) completely fixed it: Provides a continuous current path to the negative supply Enables smooth zero-crossing transitions Result: much cleaner waveform with minimal distortion Key Takeaways LM358/LM324 require careful output loading considerations in bipolar configurations Pull-down resistor to negative rail (typically 1kΩ-10kΩ) enables proper operation This is in the datasheets but easily overlooked in practice For true rail-to-rail with minimal distortion, consider CMOS op-amps (TLV274, MCP6004, etc.) Hope this helps someone debugging similar issues! The LM358 is a low cost and accessible op-amp great for general or educational/hobby use, but understanding its output stage limitations is key for clean signals. This came up while documenting some lab exercises, and I thought it was worth sharing since it's such a common gotcha. submitted by /u/eimtechnology [link] [comments]

The new entrant to the RA8 series adds real-time, high-precision motor control, strong security, and fast communications.

submitted by /u/Electro-nut [link] [comments]

Атмосфера творчості, креативу та натхнення на ЦКС Fest! kpi вт, 10/07/2025 - 21:37

The ANN-MB3 delivers centimeter-level positioning in a rugged and compact form factor, simplifying precision GNSS deployment across industries.

Up front: some background. The air-temperature sensor attached to my (home-brew) rain gauge became flaky. Short-term solution: fix it (done). Longer-term goal: improve it (read on).

That sensor is a standard Vishay NTC (negative temperature coefficient) thermistor: 10k at 25°C and with a beta value of 3977. In conjunction with a load resistor, it feeds a PIC microcontroller (MCU), which samples the resulting voltage (8 bits) for radio-linking back to base for processing and display. Figure 1 shows the utterly conventional circuit together with its response to temperature.

Figure 1 A basic thermistor circuit, together with its calculated response.

The load resistor’s value of 15699 Ω may seem strange, but that is the thermistor’s resistance at 15°C, the mid-point of the desired -9 to +40°C measuring range. Around every 30 seconds, the PIC strobes it for just long enough for the reading to settle.

Wow the engineering world with your unique design: Design Ideas Submission Guide

The plot shows the calculated response together with a straight line running through the two actual calibration points of 0°C (melting, crushed ice) and 30°C (comparison with a known-good thermometer). That response was calculated using the extended Steinhart–Hart equations rather than the less accurate exponential approximation. Steinhart and Hart (S-H) are to NTC thermistors as Callender and Van Dusen are to platinum resistance temperature detectors (PRTDs), modifying the exponential curve just as Callender-Van Dusen (CVD) tweaks an otherwise straight line.

The relevant Wikipedia article is, of course, informative. Still, a brief and useful guide to the S–H equations, complete with all the necessary constants, can be found on page 4 of Vishay’s relevant datasheet. Curiously, their tables of resistance versus temperature show truncated rather than rounded values, so they quote our device’s R15 as 15698 ohms rather than 15699. The S–H figure is 15698.76639545805…, give or take a few pico-ohms.

You’ll notice that Figure 1’s plot is upside down! That is deliberate, so a higher temperature shows a higher output, though the voltage actually falls. I think that’s more intuitive; you may disagree.

Matching an RTD to an NTC

That straight line, derived from the S–H values at 0 and 30°C, is the key to this idea. Making the PRTD generate a signal that matches it will avoid any major changes to the processing code, especially the calibration points, and it will also provide a much wider range with greater accuracy than an NTC. Because the voltage from the thermistor circuit is ratiometric, the PRTD must output a level that is a proportion of the supply.

To do that, we amplify the voltage developed across the PRTD, compensate for the CVD departure from linearity, and add an offset. The simplest circuit that can do all these is shown in Figure 2a.

Figure 2 Probably the simplest circuit (2a) that can give an output from a PRTD to match a thermistor’s response, with a slightly better variant (2b). These are both flawed, and the component values are not optimized. They are to show the principle, not the practice.

That simplicity leads to complications, because pretty much every component in Figure 2a interacts with every other one. It’s bad enough to design, even with ideal (simulated) parts, but final calibration could require hours of iterative frustration. Buffering the offset voltage, as shown in Figure 2b, helps, but that extra op-amp can be put to better use.

A practical circuit

If we split the circuit into two, life becomes easier. Figure 3 shows how.

Figure 3 The final, workable circuit. Amplification and offsetting are now separate, making calibration much easier.

The processor turns Q1 on to deliver power. (The previously active-high GPIO pin powering the thermistor must now be active-low to drive Q1’s gate, and that was the only code change needed.) The FDC604 has a low RDS(ON) of a few tens of milliohms, so it drops only 100 µV or so, which is insignificant, even if the measuring ADC’s reference is the Vdd rail. (Offsets within the MCU itself will probably be greater.) Because the circuit is only active for a millisecond every half minute or so, self-heating of the RTD can be ignored. Consumption was about 3 mA at 5 V or 2 mA at 3.3 V.

R1 feeds current through the RTD, producing a voltage that is amplified by A1a, whose gain can be trimmed by R5. R6 feeds back into the RTD and R1 to compensate for both CVD and the varying drive to the RTD as its resistance changes. Its value is fairly critical: 33k works well enough for our purposes, but 31k95—33k||1M0—is almost perfect, with a predicted error of way under 1 millidegree over a 100°C span—theoretically—so we’ll use that. Obviously, this is ridiculous overkill with 8-bit output sampling, but if a single extra resistor can eliminate one source of errors, it’s worth going for.

A1b now amplifies the signal further (and inverts it) and applies a trimmable offset. Its output as a fraction of the supply voltage is now directly proportional to the PRTD’s temperature. Note that the gain of this stage is preset: R7 and R8 should be selected so that their ratio is as close as possible to 3.9, though their absolute values are not critical. The result is shown in Figure 4.

Figure 4 Plotting the output against the RTD’s resistance now gives a result that is almost indistinguishable from the straight-line target, the (idealized) error corresponding to much less than 1 millidegree. This shows the performance limit for this circuit; don’t expect to match it in real life.

Modeling and plotting

A simple program (Python plus Pygame) to plot the circuit’s operation at different scales made it easy to see the effects of changing both R6 and A1a’s gain, with the error curve tilting (gain error) and bending (compensation error). That curve needs to be as straight and flat as possible.

Modeling the first section needed iteration, starting with a (notional) unit voltage feeding R1 and ~0.7 driving R6. Calculating the voltage across the PRTD and amplifying that gave the stage’s output, ready to feed back into R6 for recalculating V_RTD. (Repeating until successive results matched to eight significant figures took no more than ten iterations.) The section representing A1b was trivial: take A1a’s output and multiply by 3.9 while subtracting the offset.

As a cross-check, I put the derived values into LTspice and got almost the same results. The slight differences are probably because even simulated op-amp gain stages have finite performance, unlike multiplication signs.

The program also generated Table 1, which may prove useful. It shows the resistance of the PRTD at various temperatures (centered on 15°C) together with the output voltage referred to Vdd and given as a proportion of it. That output is also shown, scaled from 0–255 in both decimal and hex.

The long numbers the program generated have been rounded to more reasonable lengths, which, deliberately, are still more accurate than most test kits can resolve. Too many digits may be useful; too few never are.

Table 1 The PRTD’s resistance and Figure 3’s output calculated against temperature, centered on 15°C. The output is shown as decimals, both raw and rounded, and hex.

Compensating for long leads

As it stands, the circuit does not lend itself to true 3- or 4-wire compensation for the length of the leads to the RTD—unnecessary with an NTC’s multi-kΩ resistance. However, using a 4-wire Kelvin connection, where the power-feed and sensing lines are separate, should work well and reduce the cable’s effect, as shown in Figure 5. With less than a meter separating the RTD from the circuitry, I used speaker cable. (Copper’s TCR is close to that of a PRTD.)

Figure 5 Long leads to a PRTD can cause offset errors. Using a 4-wire Kelvin arrangement minimizes these. If the µC’s A–D has external reference-voltage pins, they can be driven from the circuit for (notionally) improved accuracy.

Figure 5 also shows how accuracy could be improved by driving the ADC’s reference pins from the circuit’s power rails, though this is academic for coarse sampling. It would also compensate for any voltage drop across Q1, should that be important. Q1 could then even be omitted, the circuit being powered directly from an active-high pin. That would drop the rail voltage, which wouldn’t matter if it were fed back to REF+.

This circuit is optimized for a center temperature of 25°C, as that is the point at which most thermistors are specified, with the load resistor equaling the R(25) value. Unlike the 15°-centered version in Figure 3, I’ve not built or tried it, but believe it to be clean. Its plot—error curve included—looked very close to that in Figure 4, but shifted by 10°C.

Errors, both theoretical and practical

The input offset voltage of op-amps changes with temperature and is a potential source of errors. The quoted figure for the MCP6002 is ±2 µV/°C (typ.), which is good but not insignificant. Heating the circuit by ~40°C (with a 100R resistor replacing the PRTD) gave an output shift corresponding to less than 0.05°, which is acceptable, and in line with calculations. (An old hairdryer is part of my workbench kit.) Here, the circuitry and the PRTD will both be outside, and thus at about the same temperature.

So how does it perform in reality? It’s now built and calibrated exactly as in Figure 3, but not yet installed, allowing testing with a PRTD simulator kludged up from resistors, both fixed and variable, plus switches so the resistance can be connected to either the circuit or a (well-calibrated) meter for precise adjustment. Checking at simulated temperatures from -10 to +50°C showed errors ranging from zero at -10° to -0.22° at +50° with either 3.3 V or 5 V supplies. This could be improved with extra fiddling (I suspect a slight mismatch in R7/8’s ratio; available parts had unhelpful spreads), but the errors are less than the MCU’s 8-bit resolution (~0.351 degrees/count, or ~2.85 counts/degree), so it’ll do the job it’s intended for, and do it well.

While this approach doesn’t substitute for a “proper” PRTD circuit, it does make a nice drop-in replacement for a thermistor, giving a wider measurement range with much better linearity while needing no extra processing. I hope the true experts in the field won’t find too many problems with it. BTW, “expert” derives etymologically from “stuff you’ve learned the hard way: been there, done that, worn the hair shirt”. Never trust an armchair expert unless you’re shopping for comfortable seating.

—Nick Cornford built his first crystal set at 10, and since then has designed professional audio equipment, many datacomm products, and technical security kit. He has at last retired. Mostly. Sort of.

 Related Content

• DIY RTD for a DMM
• Improved PRTD circuit is product of EDN DI teamwork
• Fake contacts, bounced to order
• Calculation of temperature from PRTD resistance

The post Dropping a PRTD into a thermistor slot—impossible? appeared first on EDN.

NUBURU Inc of Centennial, CO, USA — which was founded in 2015 and developed and previously manufactured high-power industrial blue lasers — says that its subsidiary Nuburu Defense LLC has secured a binding agreement to directly acquire Orbit S.r.l., an Italian software company specializing in operational resilience, business continuity, and crisis management for mission-critical organizations...

Its a Wayne Kerr 6425 LCR meter. submitted by /u/Wormjuice- [link] [comments]

After 3 weeks and studying two poorly written datasheets, I finally uploaded the initial release of my pure MicroPython driver for these graphical Futuba NAGP1250 vacuum fluorescent displays! I'm so nervous about releasing my own code lol, please be gentle I love this retro tech so much and wanted to be able to let other people share in my joy and wanted to make it as easy as possible for someone to get started! Girl power 💪 https://github.com/AlmightyOatmeal/MicroPython_Futaba_NAGP1250 girlswhocode #esp32 #womenintech #electronics #micropython submitted by /u/DangerousDyke [link] [comments]